Electromagnética Compatibility
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ALL RIGHTS RESERVED 0 l%l~ Applications for reproduction of this book or parts thereof, should be directed to: EMC Proceedings Editor, ETH Zentrum - IKT, 8092 Zurich, Switzerland. EMC Symposium & Exhibition, Zurich 7985 Honorary Patron: Mr. F. Locher, Berne Under the auspices of: Mr. R. Trachsel, Director-General of the Swiss PTT, Berne Sponsor: Association of Swiss Electrotechnicians (SEVIASE) Organized by: Institute for Communication Technology of the Swiss Federal Institute of Technology Zurich Cooperating: International Union of Radio Science (URSI), Convention of the National Societies of Electrical Engineers of W. Europe (EUREL), international Radio Consultative Committee (CCIR), IEEE Electromagnetic Compatibility Society, IEEE Switzerland Section, Association of Polish Electrical Engineers (SEP), Committee AE-4 on Electromagnetic Compatibility of the Society of Automotive Engineers (SAE), Information Technology Society of the SEV (ITG) Organizing Commlttee: Prof. Dr. P. Leuthold, Zurich (Symposium President); E. Diinner, Zurich (Vice-President); Prof. Dr. F. L. Stumpers, Eindhoven (Vice-President); Dr. T. DvoNk, Zurich (Organizing Chairman); Prof. Dr. R. M. Showers, Philadelphia (Technical Program Chairman); H. K. Mertel, San Diego (Workshops Program Chairman); U. Welte, Zurich (Exhibition Chairman); B. Szentkuti, Berne (Publicity Chairman); Dr. M. lanovici, Lausanne (Joint Events Chairman); R. Bandle, Zurich; R. Danieli, Zug; G. Meyer, Stafa; J. @rum, Zurich (Chairpersons, Local Arrangements); G. Georg, Allenwinden (Treasurer); Mrs. E. Danieli, Zug; Mrs. V. Szentkuti, Berne (Ladies Program). Technical Program Committee: Chairman: Prof. Dr. R. M. Showers Prof. Dr. P. Degauque, Villeneuve-d’Ascq; Dr. T. Dvorak, Zurich (Pro ceedings Editor); Prof. Dr. C. Egidi, Turin; Dr. J. J. Goedbloed, Eindhoven; Prof. Dr. S. Lundquist, Uppsala; Dr. A. D. Spaulding, Boulder; Dr. R. Sturm, Munster; Dr. A. Whitehouse, London; Prof. Dr. F. Zach. Wien. Advisory CommIttee: H. Bachmann, Noordwijk; Prof. Dr. F. E. Gardiol, Lausanne (Swiss National Committee of the URSI); Ft. Gressmann, Bruxelles (EBU); J. Hamelin, Lannion; J. S. Hill, Springfield (IEEE EMCS); G. A. Jackson, Leatherhead; R. C. Kirby, Geneva (CCIR); J. L. Moe, Fort Worth (SAE AE-4); Prof. Dr. J. J. Morf, Lausanne; W. Moron, Wroclaw (SEP); Prof. Dr. J. Neirynck, Lausanne (IEEE Switzerland Section); Prof. Dr. R. Sato, Sendai; Ch. Scherrer, Berne (BAUEM); Prof. Dr. Ft. Struzak, Wroclaw; Prof. Dr. A. Wedam, Ljubljana; Prof. Dr. R. Zwicky, Zurich. 79754985: Ten years of EMC Symposia Symposium Patrons 1975.1995: F. Lecher Drs. Ph. Leenman R. Trachsel Symposium Chairman: Prof. Dr. F. E. Borgnis (19751979) Sympostum President: Prof. Dr. P. E. Leuthold (1981-1985) Secretary Generallorganising Chairman: Dr. T. Dvorak (19751985) Program Chairman: Prof. Dr. F. L. Stumpers (1975-1981) Prof. Dr. Ft. M. Showers (1983-1985) Workshops Program Chalrman: H. K. Mertel (19751985) Sponsoring organlsatfons 19751965: Montreux Tourist Office, Netherlands National Committee of the IEC, Swiss Electrotechnlcal Association Organfslng tnstitutlons 1975.1965: Montreux Tourist Office, Netherlands National Committee of the IEC in cooperation with the Institute of High Frequency Electronics of the Federal Institute of Technology Zurich, Institute for Communication Technology of the Federal Institute of Technology Zurich Cooperating organlsations 19751985: International Union of Radio Science (URSI), Convention of the National Societies of Electrical Engineers of Western Europe (EUREL), International Radio Consultative Committee (CCIR), International Special Committee on Radio Interference (CISPR), Region 8 of the IEEE, IEEE Switzerland Section, IEEE Electromagnetic Compatibility Society, Association of Polish Electrical Engineers (SEP), Committee AE-4 of the Society of Automotive Engineers (SAE), Nachrichtentechnische Gesellschaft im Verband Deutscher Elektrotechniker (NTGIVDE), Information Technology Society of the SEV (ITG) Certificates of Acknowledgement: (for outstanding support of the Symposium) J. S. Hill (1977), H. K. Mertel (1977), J. C. Toler (1977). Prof. Dr. F. L. Stumpers (1983) Some data on past svmposia I Year Attendance Papers in Record Exhibitors Techn. excursions Workshops *15 summaries of the Session on Sequency 1975 1977 1979 1961 1963 396 108’ 18 444 106 19 507 107 23 529 102 25 829 103 29 -4 41 : : Techniques ” not included Prize Award Papers Honor Roil: Montreux 1971: 1. (ex aequo, in alphabetical order of the first author): lR. W. p. King, G. S. Smith: “Electrical field probes and their application in EMC” *V. P. Pevnitsky, L. V. Tigin: “A stochastic model of a cumulative. process of man-made radio interference and objective evaluation of srgnal distortions produced by these interferences” 2. D. A. Bull, G. A. Jackson: “Interference survey in military transport aircraft” 3. Ft. Struzak: “Electromagnetic compatibility: Urban electromagnetic environment - Facts models, trends” 4. R. Cortina, F. Demjchelis, W. Serravalli: “Anew type of 500kHzmeasuring instrument for long-term recording of radiointerferencefrom power lines” Montreux 1977: 1. (ex aequo, in alphabetical order of the authors): “A. P. Kalmakov: “Analysis of statistical characteristics of click voltages measured with a CISPR measuring set” *A. D. Spaulding: “Optimum reception in the presence of impulsive noise” (ex aequo, in alphabetical order of the authors): lR. J. Hasler: “The measurement of external immunity of domestic receivers-some problems and their solution” ‘R. G. Struzak: “CISPR auasi-oeak measurino channel with extended . . dynamic range” P. Groenveld, A. de Jong: “A simple r.f. immunity test setup” P. G. Galliano: irlmpulsive disturbances on car electric circuitry” Rotterdam 1979: 1. “D. Middleton: “Canonical non-Gaussian noise models: Their implications for measurements and for prediction of receiver performance” 2. (ex aequo, in alphabetical order of the first author): *I. L. Gallon: “EMP coupling to long cables” “J. Hamelin, B. Djebari, R. Barreau, J. Fontaine: ‘Electromagnetic field resulting from a lightning discharge, surges induced on overhead lines, mathematical model” “J. G. Tront, J. J. Whalen: “Computer-aided analysis of RF effects in operational amplifiers” 3. (ex aequo, in alphabetical order of the first author): W. Hadrian: Reduction of electromagnetic disturbances ‘in buildings caused by lightning using conductive facades” A. P. Kalmakov: “Possibilities of reduction of volume of measurements when checking the sources of clicks for compliance with CISPR limits” T. Takagi, t-t. Echigo, R. Sato: “Some characteristics of electric discharge as a noise source in EMC problems-recent studies in Japan” Zunbh 1981: 1. (ex aequo, in alphabetical order of the first author): ‘C. R. Paul: “Adequacy of low-frequency crosstalk prediction models” *F. M. Tesche, T. K. Liu: “Recent developments in electromagnetic field couolina to transmission lines” 2. lR. ‘Bersier: “Measurement of the immunity of TV receivers to AM RF fields in the 3 to 30 MHz range, including the influence of connected cables” 4. M. L. Crawford: “Options to open-field and shielded enclosure elec tromagnetic compatibility measurements” 5. M. Borsero, E. Nano: “Comparison between calculated and measured attenuation of the site recommended by IEC for radiation measurements” 6. B. Demoulin, P. Degauque, M. Cauterman: “Shielding effectiveness of braids with high optical coverage” Zurich 1983: 1. “J. J. Goedbloed, K. Riemens, A. J. Stienstra: “Increasing the RFI immunity of*amplifiers with negative feedback” 2. ‘T. G. Dalby: “Linear antenna near-field decoupling using a radial transmission line” 3. lB. Demoulin, P. Duvinage, P. Comic, P. Degauque: “Penetration through an interruption of the shield of a coaxial cable” 4. K. Bullough, A. Cotterill: “Ariel 4 observations of power-line harmonic radiation over North America and its effect on the magnetosphere” 5. L. E. Varakin: “Electromagnetic compatibility of cellular mobile communication systems with pseudo-noise signals” 6. J. J. Max, A. V. Shah: “Distributed lowpass filters for EMI filtering” * recipients of monetaryawards ___.___ .~ - Table of Contents A, AutomatedEMC measurements 1Al E.L.Bronaugh, P.A.Sikora, Electra-Metrics, Amsterdam, NY: Automated EMC measurements: An overview. 2A2D.N.Heirman, AT&T Laboratories, Holmdel, NJ: Automated immunity measurements. 3A3 J.C.van Essen, ESA-ESTEC, Noordwijk, Netherof an automated EMC lands: Instrumentation test facility for spacecraft. Issy-Les-Moulineaux, 4A4 G.Eumurian, Thomson-CSF, France: Computer-assisted control of EMP measurements on major systems. B, ESD techniques 5Bl P.Richman, A.Tasker, KeyTek Instrument Corp., Burlington, MA: ESD testing: The interface between simulator and equipment under test. 6B2 M.Mardiguian, D.R.J.White, Don White Consultants, Inc., Gainesville, VA: Recent develop: ments in the understanding of coupling paths of ESD through a metallic cabinet. 7B3 L.Inzoli, Honeywell ISI, Milano, Italy: -ESD susceptibility and radiated emissions of EDP peripheral printers. 8B4 B.Daout, H.Ryser, Hasler Ltd., Berne, Switzerland: Fast discharge mode in ESD-testing. C, TriggeredlightningEMP WI I@2 UC3 DC4 H.Kikuchi, Nihon University, Tokyo, Japan: A new model of triggered lightning. The St.Privat d'Allier Research Group, France: Applications of triggered lightning in France: Possibilitiesand limitations. A.S.Podgorski, NRC, Ottawa, Canada; J.A.Landt, Los Alamos National Laboratory, NM: Numerical analysis of the lightning - CN tower interaction. T.Takeuti, M.Nakano, Z.-I.Kawasaki, N.Takagi, Nagoya University, Toyokawa, Japan: Electromagnetic fields on the ground due to lightning strokes triggered with rockets and a tall chimney. D, EMC measurements UDl J.D.Gavenda, University of Texas; J.H.Davis, IBM Corp., Austin, TX: Electromagnetic wave propagation in a semi-anechoic chamber. 14D2M.Kanda, NBS, Boulder, CO: A methodology for evaluating microwave anechoic chamber measurements. 15D3 S.C.Kashyap, NRC, Ottawa, Canada: Field distortions in a TEM cell. 16D4J.H.Davis, W.C.Cockerill, IBM Corp., Austin, TX: Chamber quality assessment. 17D5S.Linkwitz, Hewlett-Packard Co., Santa Rosa, CA: Discriminating between narrowband and broadband EM1 using a spectrum analyser. 18W U.Raicu, G.U.Sorger, Eaton Corp., Sunnyvale, CA: Broadband YIG-tuned preselector for VHF and UHF. 1gD7 G.K.Boronichev, LONIIR, Leningrad, USSR: Measurement of the immunity of broadcast receivers according to the CISPR method and the difficulties encountered. Technical University of Wroc2oD8 T.W.Wigckowski, law, Poland: On the measurement of EM power density using a double loaded loop antenna. E. Printedcircuit board EMC ZEl C.R.Paul, University of Kentucky, Lexington, KY: Printed circuit board EMC. 2232B.Danker, N.V.Philips, Eindhoven, Netherlands: New measures to decrease radiation from printed circuit boards. 23~3D.R.Bush, IBM Corp., Lexington, KY: Radiated emissions of printed circuit board clock circuits. 24~4H.W.Ott, AT&T Bell Laboratories, Whippany, NJ: Controlling EM1 by proper printed wiring board layout. 25~5R.F.German, IBM Corp., Boulder, CO: Use of a ground grid to reduce printed circuit board radiation. 26~6J.W.E.Jones, Portsmouth Polytechnic, England: Achieving compatibility in inter-unit wiring. 2737J.P.Charles, CNET, Issy-Les-Moulineaux, France: Electromagnetic interference control in logic circuits. F, Lightningelectromagnetic pulse 289 C.D.Weidman, E.P.Krider, University of Arizona, Tucson, AZ: Lightning radiation fields. 2%'2F.Heidler, Hochschule der Bundeswehr Muenthen, Neubiberg, GFR: Traveling current source model for LEMP calculation. 30-3C.Weidman, J.Hamelin, M.Le Boulch, CNET, Lannion, France: Radiation characteristics, emission mechanisms and phenomenology of lightning. 31F4M.W.Wik, Defence Material Administration, Stockholm, Sweden: Double exponential pulse models for comparison of lightning, nuclear and electrostatic discharge spectra. 32~5R.L.Gardner, L.Baker, MRC, Albuquerque; C.E. Baum, D.J.Andersh, Kirtland AFB, NM: ~Comparisen of lightning with public domain HEMP waveforms on the surface of an aircraft. 33F6D.Jaeger, R.Rode, MBB GmbH, Ottobrunn, GFR: NEMP and lightning protection requirements for modern aircraft equipment. 34F7F.Pigler, Siemens AG, Erlangen; P.Kronauer, BBC, Mannheim; R.Terzer, KWU, Erlangen, GFR: Prediction of lightning-induced interference voltages on the basis of measurements taken in similar installations. 35F8H.Schiippler, D.Ristau, University of Transport, Dresden; H.Lorke, IPF, Berlin, GDR: Impulse current and voltage propagation in underground telecommunication cables. G, EM wave interactionwith biological systems 3&l Q.Chen, R.C.Huang, B.C.Pan, CARIS, Beijing, China: The hazard of electromagnetic radiation and discussion of safety thresholds. 37~2 T.S.Tenforde, C.T.Gaffey, M.S.Raybourn, University of California, Berkeley, CA: _... Influ_-. ence of stationary magnetic fields on ionic conduction processes in biological systems. Centre, N.Dekleva, D.Vujnovid, Clin.Hospital Zemun; B.Beleslin, Medical Faculty; V.Majid, Electrotechnical Faculty, Belgrade, YUgOSlavia: Magnetostimulation - A method for reestablishment of antibiotic bactericidal action. CNRS, Thiais, France: Specific 3964 A.J.Berteaud, mechanisms of microwave power dissipation in living tissues. 40G5 R.G.Olsen, Naval Aerospace Medical Res.Laboratory, Pensacola, FL: Measurement of specific absorption rate in a full-size man model near a 10.67-m monopole antenna/ground plane system at 2.101 MHz. 41G6G.d'Ambrosio, A.Scaglione, F.De Martino, R. Pennarola, University of Naples, Italy: Ku_ band radiation effects on the eye. 42~7 D.W.Griffin, N.Davias, University of Adelaide, Australia: Wideband evaluation of microwave intensity near the eyes with scattering structures present such as safety spectacles. 3803 H, Statisticalaspectsof noise and limits 43HlA.de Jong, Dr.Neher Laboratories PTT, Leidschendam, Netherlands: Statistical aspects of noise and limits. Department of Trade and InA.C.D.Whitehouse, dustry, London, England: Radio interference - The probability problem. 45H3B.Audone, R.Cazzola, G.Barale, Aeritalia, Torino, Italy: Statistical evaluation of the EMC safety margin at system level. 46H4R.Bersier, Swiss PTT, Berne, Switzerland: -The state of art of TV receiver immunity and recommendations for appropriate construction deduced from test statistics. 47H5A.P.Kalmakov, LONIIR, Leningrad, USSR: Probability distributions of effective voltages of man-made radio interference and their use for the calculation of limits. 48H6Q.Chen, Y.C.Zhu, CARIS, Beijing, China: The application and development of EMC in China. I, EM Phenomenain Power transmissionand distribution 4911 H.-J.Haubrich, VEW AG, Dortmund, GFR: New ways for interference computation and MonteCarlo-optimization to guarantee the compatibility of inductively coupled line systems. 5012 W.MachczyAski, Polytechnic of Poznan, Poland: Potentials and currents along an earthed buried cable exposed to electromagnetic effects of a power line under fault condition. 5113J.L.ter Haseborq, H.Trinks, Technical University Hamburg-Harburg; R.Sturm, NBC Defence Research and Development Institute, Munster, GFR: Coupling and propagation of transient currents on multiconductor transmission lines. 5214F.Paladian, J.P.Plumey, D.Roubertou, J.Fontaine, University of Clermont-Ferrand, France: Response of a single-conductor overhead wire illuminated by an inhomogeneous plane wave. 5315F.Maumy, B.Jecko, O.Dafif, University of Limoges, France: Time domain scattering by thin wire structures above a homogeneous ground. 916 %I1 SchwaA.Strnad, H.RGhsler, Energie-Versorgung ben, Stuttgart, GFR: Noise sources and interference values in high voltage substations. T.Yoshino, I.Tomizawa, University Of Electrocommunication, Tokyo, Japan: Balloon and satellite observation of power line radiation over northern Europe. J, Computerprogramsfor the EMC engineer SJI J.K.Breakall, G.J.Burke, E.K.Miller, Lawrence Livermore National Laboratory, Livermore, CA: The numerical electromagnetic code (NEC). 5752 D.J.Bem, J.Janiszewski, R.Zielidski, Technical University of Wroclaw, Poland: Computer. aided analysis of electromagnetic compatibllity in VHF-FM broadcasting networks. 58J.3 A.Farrar, NTIA, Annapolis, MD: Computer models for determination of satellite powerflux-density limits. University of Tsukuba, Ibaraki, 5qJ4 K.Hirasawa, -- Japan: Computer programs for calculating bounds of interference between arbitrarily shaped wire antennas. aJ5 G.Azrak, Merlin-Gerin, Grenoble; Ph.Auriol, Ecole Centrale de Lyon, Ecully, France: Numecompati~rical simulation of electromagnetic bility in time domain. 61~6W.Krzysztofik, Technical University of Wroclaw, Poland: Electromagnetic wire scattering of thin cylindrical antennas loaded by nonlinear impedances. K, EMI in microelectronics 62~1J.J.Whalen, SUNY at Buffalo, Amherst, NY: Determining EM1 in microelectronics - A review of the past decade. 63~2J.G.Tront, Virginia Polytechnic and State University, Blacksburg, VA: Comparison of the RF1 susceptibility of several typical IC pin drivers/receivers. SUNY at Buffalo, Am64~3 Y.-H.Sutu, J.J.Whalen, herst, NY: Demodulation RF1 in inverting and non-inverting operational amplifier circuits. L. Nuclearelectromasnetic Pulse imoact 65~1O.Dafif, C.Bardet, E.Jecko, University of Limoqes, France: Transient field distribution in a transmission line simulator. 66~2H.-D.Briins, D.KBniqstein, Hochschule der Bundeswehr, Hamburg, GFR: Calculation and measurements of transient electromagnetic fields in EMP simulators. 67~3T.Karlsson, G.Unden, M.Gylemo, National Defence Research Institute, Linkoeping, Sweden: EMP simulation by pulse injection. 68~4 M.E.Gruchalla, A.J.Bonham, J.Gibson, P.G. Johnson, EG&G WASC Inc., Albuquerque, NM: A portable programmable pulser and high-speed, log-weighted peak-level recorder for direct.___ drive testing. 69L5C.E.Baum, AWFL, Kirtland AFB, NM: Black box bounds. 70L6 P.B.Johns, University of Nottingham; A.Mallik, Kimberley Communications Consultants, Nottingham, England: EMP response of aircraft structures using transmission-line modelling. 71=7I.L.Gallon, AWRE, Aldermaston, England: Radiation damping in finite cylinders. 72L8A.Caron, B.Djebari, A.Zeddam, CNET, Lannion, France; Ph.Blech, Y.Dijamatovic, M.Ianovici, EPFL, Lausanne, Switzerland: Validation of EMP calculation methods using the response of an aerial cable to a lightning stroke. M. Power and data line transients EMI 74~2 75M3 76M4 77M5 & W.T.Rhoades, Xerox Corp., El Segundo, CA: Characteristics of unusual power main transients. F.D.Martzloff, GEC Comp., Schenectady, NY: The development of an IEEE guide on surge testing for equipment connected to low-voltage AC power circuits. P.Richman, KeyTek Instrument Corp., Burlington, MA: Changes to classic surge-test waves required by back-filters used for testing powered equipment. V.Scuka, Uppsala University, Sweden: Performance deterioration of metal oxide varistors by current surges. M.Tetreault, Digital Equipment Corp., Stow, MA; F.D.Martzloff, GEC Comp., Schenectady, NY: Characterization of disturbing transient waveforms on computer data communication lines. Statisticaltheory of EMC 78Nl D.Middleton, New York, NY: Threshold signal and parameter estimation in non-Gaussian EMC environments. 7gN2 A.D.Spaulding, NTIA, Boulder, CO: Locally optimum and sub-optimum detector performance in non-Gaussian "broadband" and "narrowband interference environments. 8fjN3N.N.Buga, V.Y.Kontorovich, Electrotechnical Institute of Communications; Y.V.Polozok, LONIIR, Leningrad, USSR: Electromagnetic environment control on the basis of system models with random structure. 0, Spread spectrumand mom communications 8101 P.M.Hopkins, D.N.Cravey, Lockheed Co., Inc., Houston, TX: Spread spectrum communications - Interference considerations. 8202 K.Dostert, University of Kaiserslautern, GFR: EMC problems in data transmission over indoor power lines using spread spectrum techniques. 8303 L.E.Varakin, All-Union Telecommunication Institute by Correspondence, Moscow, USSR: The efficiency of the cellular spread spectrum radiotelephone. H.Ochsner, Federal Institute of Technology, 8404 Zurich, Switzerland: Comparison of spectrum efficiency of CDMA and FDMA mobile radio systems. G.K.Chan, Department of Communications, Ot8505 tawa, Canada: Interference analysis of a land mobile cellular radio system. 8606 K.Fisher, Department of Trade and Industry, London, England: Planning of television band III for use by mobile services. 8707 B.BeriE, Federal Radiocommunication Direction, Belgrade, Yugoslavia: Comparison of field strength measurements and computer prediction in land mobile service. 8808 A.Golas, Telecommunication Research Centre, New Delhi, India: Compatibility of TV and UHF communications antennas mounted on the same tower. 899 S.Satyamurthy, Combat Vehicles R&D Establishment, Madras, India: Design of compatible equipment for land mobile vehicles. P, Shieldingand cable coupling WPl S.R.Ramasamy, Defence Electronics Research Laboratory, Hyderabad; S.Mahapatra, Indian Institute of Technology, Bombay, India: Attenuation of electromagnetic radiation from microwave ovens utilizing corrugated metallic surface combined with magnetic resistive sheets and absorbers. 91P.2W.Hadrian, Technical University of Vienna, Austria: Low-frequency magnetic shielding effectiveness of steel-reinforced concrete platforms. 92P3 B.L.Michielsen, Philips Research Laboratories, Eindhoven, Netherlands: A new approach to electromagnetic shielding. 93p4V.A.Morozov, N.V.Rodionova, USSR Academy of Sciences, Moscow, USSR: Field nonuniformity reduction inside a spherical magnetic shield. 94P5 H.Rahman, St.Louis University, Cahokia, IL; J.Perini, Syracuse University, NY: EMP enclosure penetration and cable coupling. !%P6 B.Demoulin, P.Duvinage, P.Degauque, Lille University, Villeneuve d'Ascq, France: Measurements of transfer parameters of shielded cables at frequencies above 100 MHz. 96P7 K.H.Gonschorek, Siemens AG, Erlangen, GFR: Magnetic stray fields of twisted multicore cables and their coupling to twisted and nontwisted two-wire lines. Q, Power electronics 97Ql M.Di Stefano, Italian Railways, Roma; G.L. Solbiati, SIRTI S.p.A., Milano, Italy: -Project of a railway electrification from the EMC point of view. 98Q2 H.Kunkel, M.Lutz, O.Frey, High Voltage Test Systems, Basel, Switzerland: Coupling and filtering possibilities of transients during EMC tests. ggQ3 F.C.Zach, Technical University of Vienna, Austria: A new pulse width modulation control for.line commutated converters minimizing the mains hacmonics content. 10op4 J.Sack, H.Schmeer, Hochschule der Bundeswehr Muenchen, Neubiberg, GFR: Computer-aided analysis of the RF1 voltage generation by small commutator motors. 101Q5 J.M.Firth, NRC, Ottawa, Canada: Control and reduction of spurious emissions from small DC to DC power converters. 1@Q6 B.Brdndli, J.Bertuchoz, R.Steck, NC-Laboratory, Spies, Switzerland: High current fast pulse measurement with a Rogowski coil. 103Q7 V.Nikiforova, All-Union Research Institute of Energetics, Moscow, USSR: Electromagnetic compatibility of electrical equipment in power and industrial supply systems. S, Systems EMC and protectivemeasures u@l H.Cichofi,H.Trzaska, SARU Region 1 EMC Working Group, Poland: Selective interference in home entertainment electronic devices. US2 I.Oka, K.Ishida, I.Endo, University of Electro-Communications, Tokyo, Japan: Co-channel interference in an on-board processing satellite. R, Key Problemsof spectrumuse 112.53S.Yamazaki, H.Kuronuma, NHK Science & Technil@Rl K.Olms, FTZ, Darmstadt, GFR: Radio frequency cal Laboratories; Y.Noguchi, Nippon Electric spectrum management. Company, Tokyo, Japan: Relation between APD/ l&R2 H.J.Weiss, COMSAT, Washington, DC: The big CRD of automobile ignition noise and resulsqueeze - A selective look at ORB-85/88. tant TV picture degradation. l&R3 A.H.Wojnar, Warsaw Academy of Technology, PoUS4 K.Uchimura, T.Aida, Kumamoto University; T. land: Deformable lattices for efficient freTakagi, Tohoku University, Sendai, Japan: quency management. Electromagnetic radiation caused by silver 107R4 R.Sandell, BBC Research Department, Tadworth, palladium alloy contact switching. England: The prediction of field strength in u4S5 W.van Eck, J.T.A.Neessen, P.J.M.Rijsdijk, Dr. the frequency range 30-1000 MHz and its inNeher Laboratories PTT, Leidschendam, Netherfluence on spectrum management. lands: On the characteristics of the electro1@~5 G.A.De Couvreur, M.C.Delfour, Department of magnetic field generated by video display Communications, Ottawa, Canada: Optimum freunits. quency assignment strategies for radio celluu5S6 W.Biichler,Meteolabor AG, Wetzikon, Switzerlar land: Overvoltage protection circuits. 1@R6 P.Vaccani, Department of Communications, Ot116~7 M.A.Bykhovsky, G.G.Gurianov, Ministry of Tetawa, Canada: A second generation mobile speclecommunications, Moscow, USSR: Iterative trum monitoring system. interference simulator for the division of two FM signals. -c-m- Authors 4 Aida T. Andersh D.J. Audone B. Auriol Ph. Azrak G. 113s4 32F5 4583 6055 6OJ5 B - Baker L. Barale G. Bard& C. Baum C.E. Beleslin B. Bern D.J. BeriB B. Bersier R. Berteaud A.J. Bertuchoz J. Blech Ph. Bonham A.J. Boronichev G.K. Braendli B. Breaknll J.K. Bronaugh E.L. Bruens H.-D. Buechler W. Buga N.N. Burke G.J. Bush D.R. Bykhovsky M.A. 32F5 4583 65Ll 3235, 69L5 3863 57J2 8707 4684 39G4 10296 72L8 68L4 19D7 102Q6 56Jl 1Al 66L2 115S6 8ON3 5651 2333 11657 c Caron A. cazzo1.3 R. Chan G.K. Charles J.P. Chen Q. Cichofi H. Cockerill W.C. Cravey D.N. 72L8 4583 8505 2737 36G1, 4886 llOS1 16D4 8101 D - Dafif 0. d'Ambrosio G. Danker 8. Daout B. Davias N. Davis J.H. De Couvreur G.A. Degauque P. de Jong A. Dekleva N. Delfour M.C. De Martin0 F. Demoulin B. Dijamatovic Y. Djebari B. Di Stefano M. Dostert K. Duvinage P. 5315, 65Ll 41G6 22E2 SB4 42G7 13D1, 1604 108R5 95P6 43Hl 38G3 108RS 41G6 95P6 72LB 72L8 97Ql 8202 95P6 H Gruchalla M.E. Gurianov G.G. Gylemo M. 68L4 11657 6n3 Hadrian W. Hamelin J. Haubrich H.-J. Heidler F. Heirman D.N. Hirasawa K. Hopkins P.M. Huang R.C. 91P2 30F3 4911 29F2 2A2 59J4 8101 36Gl 1 Ianovici M. Inzoli L. Ishida K. J Jaeger D. Janiszewski J. Jecko B. Johns P.B. Johnson P.G. Jones J.W.E. I(_Kalmakov A.P. Kanda M. Karlsson T. Kashyap S.C. Kawasaki Z.-I. Kikuchi H. Kijnigstein D. Krider E.P. Kontorovich V.Y. Kronauer P. Krzysztofik W. Kunkel H. Kuronuma H. c Landt J.A. Le Boulch M. Linkwitz S. Lorke H. Lutz M. fl Machczydski W. Mahapatra 6. Majid V. Mallik A. Mardiguian M. Martzloff F.D. Maumy F. Michielsen B.L. Middletoh D. Miller E.K. Morozov V.A. E Endo I. Eumurian G. lllS2 4A4 N Nakano M. - Neessen J.T.A. Nikiforova V.N. Noguchi Y. E Farrar A. Firth J.M. Fisher K. Fontaine J. Frey 0. 58~3 lOlQ5 8606 5214 9BQ2 0 Ochsner H. _. Oka I. 0lms K. Olsen R.G. Ott H.W. 5 Gaffey C.T. Gallon I.L. Gardner R.L. Gavenda J.D. German R.F. Gibson J. Galas A. Gonschorek K.H. Griffin D.W. 37G2 71L-l 32F5 13Dl 2535 68L4 8808 96P7 42G7 Index fl Paladian F. Pan B.C. Paul C.R. Pennarola R. Perini J. Pigler F. Plumey J.P. Polozok Y.V. Podgorski A.S. 72L8 783 11152 33F6 57J2 5315, 65Ll 7OL6 68L4 26~6 47H5 14D2 6n3 15D3 12C4 9Cl 66L2 28Fl 8ON3 34F7 6156 98Q2 112S3 llC3 30F3 17D5 3538 98Q2 5012 9OPl 3aG3 7OL6 6~2 74M2, 77M5 5315 92P3 78Nl 56Jl 93P4 12C4 11455 103Q7 112S3 8404 lllS2 104Rl 40G5 2434 5214 36Gl 21El 41G6 94P5 34F7 5214 8ON3 llC3 B R&man H. Raicu D. Pamasamy S.R. Raybourn M.S. Rhoades W.T. Richman P. Rijsdijk P.J.M. Ristau D. Rode R. Rodionova N.V. Roehsler H. Roubertou D. Ryser Ii. 94P5. 18D6 9OPl 3702 73Ml 5B1, 75M3 11455 35FS 33F6 93P4 5416 5214 a34 5 Sack J. Sandell R. Satyamurthy 8. Scaylione A. Scbmeer H. Schiippler H. Scuka V. Sikora P.A. Solbiati G.L. Sager G.U. Spaulding A.D. Steck R. St.Privat d'Allier Research Group Strnad A. Sturm R. Sutu Y.-H. lOOQ4 107R4 8909 4166 lOOQ4 35FS 76M4 1Al 97Ql 18D6 79N2 102Q6 1 Takagi N. Takagi T. Takeuti T. Taker A. Tenforde T.S. ter Haseborg J.L. Terzer R. Tetreault M. Tomizawa I. Trinks H. Tront J.G. Trzaska H. 12C4 11354 12C4 5Bl 37G2 5113 34F7 77M5 5517 5113 6332 llOS1 u Uchimura K. Und&n G. 113s4 6n3 V Vaccani P. van Eck W. van Essen J.C. Varakin L.E. Vujnovid D. 109R6 11455 3A3 8303 38G3 w Weidman C.D. Weiss H.J. Whalen J.J. White D.R.J. Whitehouse A.C.D. Wi?ckowski T.W. Wik M.W. Wojnar A.H. v Yamazaki S. Yoshino T. z Zach F.C. Zeddam A. Zhu Y.C. Zielihski R. loC2 5416 5113 64K3 28F1, 3OF3 105R2 62KI, 64K3 6~2 44H.2 20D8 31F4 106R3 11253 5517 9993 72L8 48H6 57J2 Scientific Contributions - 1 Al 1 - AUTOMATED EMC MEASUREMENTS: AN OVERVIEW Edwin L. Bronaugh and Paul A. Sikora Electra-Metrics 100 Church Street Amsterdam, New York 12010 USA Abstract This paper looks at the history of automated EMC measurements and the current technology. It discusses the scope of this session. A philosophical discussion is included to lead to understanding the strengths and weaknesses of current technology and needs for future development. A present-day computer-controlled interference emissions measuring system is described. Introduction Background The desire for automated EMC measurements found its inception in the decade of the 1950's with the greatest incentive arising, perhaps, from the plethora of measurements mandated by military EMC standards on military communications and electronics equipment. The problem most pressing at the time could be summarized as too many measurements to be made resulting in too much data to analyze all in too short a time. From this apparent need arose mechanical attachments for the manual radio noise meters of the day to tune them automatically over their available tuning ranges while driving the X-axis of an X-Y plotter with a voltage proportional to the position of the mechanical tuning mechanism and the Y-axis with the envelope voltage from the indicating instrument drive circuitry. Although these "automatic" instruments were crude and frequently inaccurate, they provided the data much faster, more reliably, and in a more usable graphical form than could be provided by a human operator tuning, reading an indicator, and writing down the data on a pointby-point basis. For several years, EMC instrument manufacturers worked to improve upon this early swept tuned instrument by providing electronic tuning, more responsive detector functions, large dynamic measurement range by use of AGC or logarithmic amplifiers, and untuned wide band antennas and transducers. To this day such instruments are still widely used to make measurements in accordance with MIL-STD-461/ 462 [4, 51 and other military standards, During this same time, spectrum analyzers or panoramic receivers were being developed for somewhat different purposes, but would eventually come to be used for some EMC measure- ments. Then came the era of the computer, and EMC engineers and instrument manufacturers saw advantages to the use of computers to control the EMC test instruments. The computer could operate the test instruments; record data; apply antenna, transducer, cable loss, instrument calibration, and other correction and conversion factors to the data; and plot this reduced data on multi-decade plots for ease in comparing the performance of equipment under test with the limits in the technical standards. Many such systems for measuring interference emissions are in use today. While much automation has been achievedwith interference emissions tests, automation of interference immunity (susceptibility) tests has lagged far behind. One of the many reasons for this has been the more complicated nature of immunity tests. Purposes and Objectives This paper has two purposes. One is to introduce this session on automated EMC measurements by giving an overview of the session, and by discussing the philosophy of automated EMC measurements. The other purpose is to present some details on an automated radio noise (EMI) measuring system incorporating both self- and computer-controlled test capabilities. The objectives are to bring out some of the strengths and weaknesses of automated EMC tests, and to stimulate thinking towards continuing improvement in EMC measurements. The scope of this session is to address the issues associated with the use of computercontrolled or self-controlled automatic and semi-automatic test equipment and techniques to make EMC measurements* Both halves of the EMC test question will be addressed, i.e., both emissions (interference) and immunity (susceptibility) measurements. Some of the automated EMC measurement issues to be raised and discussed are: 1. Emissions and immunity testing for regulatory compliance versus testing for engineering and development; 2. The effects of the test equipment scan rate, the statistics of the radio noise or disturbance being measured or simulated, a mixture of signals and noise, and the characteristics of sources on EMC measurements; 3. Automation of EMC measurements as a tool to - correct deficiencies in present manualmeasurement techniques; 4. EMC measurements for meeting military requirements contrasted with those for commercial requirements (import licensing, type approval, etc.); and 5. Special characteristics of test instrumentation for automated EMC measurements versus that traditionally used in manual EMC measurements. Automated EMC Measurements Before we can improve upon the automatedEMC measurement instrument and extend its use to non-military EMC measurements, we must ask and answer a number of questions. Some of these questions are: 1. Why make automated EMC measurements? 2. What needs to be measured? 3. What are the strengths and weaknesses of the present technology? 4. What goals should be set for advancingautomated EMC measurement technology? To answer some of these questions and see the need to ask others, we must understand the sometimes conflicting requirements of EMC measurement standards and regulations. What do the technical standards of such bodies as IEC, CISPR, ANSI, ISO, VDE, FCC, CSA, JASO, SEV, and many others, and the military establishments of several countries have in common and where do they differ? Can an automated system be made to adequately deal with the differences? Should tasks that a thinking human operator can do easily be automated for an unthinking computer controller to do poorly, e.g., click measurements [a]? Why Make Automated EMC Measurements? This question was basically answered in the introduction, and the reasons are yet with us these days. Even though we have achieved some degree of automation in EMC measurements, the growth of the use of electronics with its concomitant growth of EMI causes the problem mentioned earlier of "too many measurements, too much data, and too little time," to be with us continually. This places us in a dilemma of needing either more and better automation of EMC measurements or fewer EMC measurements to make. The latter choice would tend to imply less or poorer control of radio noise andelectromagnetic interference, and, thus, a worsening lack of compatibility among our uses of the electromagnetic spectrum and electrical and electronic appliances and equipment. Even now, some regulatory agencies are reducing the amount of testing for compliance with their regulations in attempts to achieve a more acceptable balance between the amount of testing needed and the degree of EMC obtained,with the economics of both issues being a major consideration, What Needs to be Measured? A complete answer to this question is beyond the scope of this paper; however, an outlineof the answer would obviously include both interference emissions from equipment and interference immunity (susceptibility) of equipment. Also, both conducted and radiated interference emissions and immunity measurements would be included. To get into a more detailed analysis of what to measure, one must study in detail 2 - the various EMC measurement standards and regulations along with the current and predicted EMC problems in the geographical area of interest. The fact that geographical areas are an important factor is obvious if the interference regulations of high population density, high technology areas are compared with those of low population density, high technology areas. Strengths and Weaknesses of Automated EMC Measurements The present technology for automated EMC measurements uses desk-top calculators and small computers to operate the test instruments, record and reduce data, provide calibration, plot results, and even write test reports. At its present state, the technology has both advantages and disadvantages. Some of the advantages are summarized by Mr. D.N. Heirman of AT&T in his paper [l] in this session: "Automation of EMC Testing should be viewed as an engineering tool and not as a replacement for the engineer who must determine compliance with either regulatory or corporate EMC criteria. As a tool, automation if implemented properly decreases the likelihood of measurement error due to operator inattention, test instrumentation misadjustments, and inability to recreate all the test conditions on a repeatable basis. The cost of automation must also be weighed against the increased test time normally associated with manual operations." The advantages of these automated test systems seem to be abundant, but they also have serious disadvantages. Automation of EMC tests now often provides us with much more incorrect data faster. It is difficult to make the computer think and understand what is being measured, while the human operator who can think and understand can't make measurements as fast and tends to record data incorrectly or lose it entirely. However, a major reason for the incorrectness of the automated data lies in the typical understanding of the state-of-the-art many years ago. Many an EMC engineer is so happy to have the measurements done quickly with less labor that he or she has forgotten that many measurement errorproducing compromises were the state-of-theart years ago and are still present andaffect the correctness of the data taken by modern instruments. An example of this is in field strength measurements. Most EMC field strength measurements are made under conditions in which the electromagnetic fields to be measured are not homogeneous, but the antennas used to make the measurements are calibrated for and operate properly only in homogeneous, planewave fields which might be found in free space many wavelengths from their sources. Mixed signals and noise can pose particularly difficult problems. An example of this may be seen in testing a vehicle for ignition noise emanations. The vehicle is to be usedin the vicinity of sensitive receivers for long wave, medium wave, short wave, etc. communications, broadcasting, and navigation; thus, its ignition noise must meet stringent limits from 10 kHz to 1 GHz. The vehicle is large and the testing organization has no large shielded chamber in which to test it, so it must be tested outside. The outside environment - contains much noise and many narrowband signals throughout the required test frequency range, and most of these signals are so large that they produce indications in the EMI analyzer or radio noise meter far above the limit specified for the ignition noise emanations from the vehicle under test! Current EMC Instrumentation Technology The radio noise meter characteristics [2, 31 are the primary factor that determines if the ignition noise in the above example can be measured throughout the range of frequencies from 10 kHz to 1000 MHz. The regulatory requirements [4, 5, 6, 71 are secondary factors in the accurate and successful measurement of EMI in such a non-ideal real-world measurement situation. A typical simulation of the array of input signals and noise which the radio noise meter must resolve in the example above may consist of: 1) The signal from an impulse generator set to produce a level of 52 dB(uV/MHz) which simulates the vehicle ignition noise; 2) A pulse generator operating at 50 kHz producing a pulse amplitude of 0.0025 ~VS [68 dB(uV/MBz)] which simulates low frequency industrial noise in the vicinity of the test site; and 3) Two cw signal generators set to produce signals at 22 kHz and 8 MHz at levels of 64 dB(pV) and 49 dB(llV), respectively, to simulate two of the many narrowband communications and broadcast signals also in the ambient of the test site. A thinking, well-trained and experienced human operator using an ordinary radio noise mater would have an exceedingly difficult time resolving this spectrum of noise to determine correctly the level of the impulse generator, but this appears to be an almost impossibleto-solve problem using a computer-controlled radio noise meter unless it and the control computer software have capabilities that exceed those usually found in "standard" EMI analyzers or radio noise meters and controllers. In the 10 kBz to 150 kHz frequency range, the standard CISPR radio noise meter [2] has a bandwidth of 200 Hz and the standard ANSI radio noise meter [3] has bandwidths of 200 Hz, 1 kHz, and 10 kHz. In a 200 Hz bandwidth the impulse generator produces a level of -22 dB(!.N), the pulse generator produces a level of -6 dB(pV), and the 22 kHz cw generator produces a level of 64 dB(UV). In a 1 kHz bandwidth these levels become -8 dB(lN), +8 dB(!Jv) , and 64 dB(pV), respectively. In a 10 kHz bandwidth, the levels become +12 dB(uV), +28 dB()N), and 64 dB(BV), respectively. The simulated ignition noise (the impulse generator) which must be measured is far below the interfering signals and may be below the impulse sensitivity of the radio noise meter in a 200 Hz bandwidth. As can be seen from the above data, when the bandwidth of the radio noise meter is made larger to bring the simulated ignition noise UP to a level where it can be easily measured, the bandwidth is so wide that the 22 kHz narrowband signal begins to override the simulated ignition noise in the skirts of the radio noise meter selectivity characteristic. This effectively prevents the detector in the radio noise meter from properly responding to the simulated ignition noise, It may be seen that theproblem of relative noise levels continues on above 50 kHz, 3 - 1 Ad A well trained, experienced human operator using visual techniques with an oscilloscoPe on the radio noise meter output may be able to make some satisfactory measurements, given enough time. A computer-controlled analyzer would need to be extremely sophisticated to do as well as the human operator. perhaps the best that could be done by an automated system, would be to determine that no satisfactory broadband noise measurement could be made in this frequency range under these conditions, and so inform the operator. Because of problems such as this, the US Air Force has seen fit to issue an application note [61 recommending that "official" measurements be made in one bandwidth and compared against one limit no matter what the nature of the EMI, broadband or narrowband. The United Kingdom is in the process of issuing regulations to this effect [7]. Both of these documents assume that measurements can always be made in a low ambient noise environment, such as a shielded enclosure, although this is often not possible. In the current technology, CISPR and ANSI instruments are specified in such a manner as to imply that manual EM1 measurements are to be made. At the same time, the military presumes 153 that some form of automated measurements will be made, and test laboratories performing EMI measurements to comply with military standards are generally making automated measurements. Also, automation has begun to pervade EMI measurements made to comply with standards and regulations, such as those of the VDE [8] and FCC [9], covering consumer electronics equipment. The above discussion applies only to the measurement of EMI, but similar instrumentation problems exist in making interference immunity (susceptibility) measurements. From one viewpoint, the worldwide community of EMC scientists and engineers is better off with respect to making immunity measurements since few regulations exist covering these measurements. This allows those who wish to make immunity measurements much freedom to develop instrumentation and methods that are timely and appropriate. Mr. Heirman demonstrates this in his Paper CU. This does not mean that automated immunity measurements are intrinsically any easier to make or more reliable than automated emissions measurements. Immunity measurements will be addressed by other papers in this session. An Automated Measurement and Analysis System The problem posed above wherein several different signals and noises are superposed was investigated further with the objective of finding a way to automate the measurements and yet obtain valid results. First, the needed attributes of the system are discussed, then ways one might manually measure the various signals and noises are investigated, and finally a method combining hardware and software is realized. The discussion applies to the 10 kBz to 150 kHz frequency range, but similar problems exist in, and similar techniques can be applied to, other frequency ranges. In order to insure that impulsive and cw signals are properly measured, the automated system must be able to make several decisions without manual intervention by the operator. First the system must be able to identify all -4- cw signals. This can be done with the use of a discriminator which recognizes these signals when they are encountered. This is relatively easy to do by monitoring the FM video output of the receiver for a D.C. shift in the output level. Once the presence of the cw signal is determined, a more difficult decision must be made: Is there a significant impulse level superposed on the cw signal? Since all NIL-STD type measurements must be made with a peak detector, the level of the cw signal read includes any additive impulse level; the problem is to isolate the impulsive signal and measure its level. Because of the logarithmic scaling of the radio noise meter output level, a high level cw signal can almost entirely mask an impulsive signal. By way of example, consider a situation where one finds a narrowband signal present at 15 kHz at a relatively high level, 60 dB(uV). A typical source of such a narrowband signal would be switching regulated power supply operating at a switching frequency of 15 kHz. The narrowband signal is one line, the fundamental, of the spectrum of many harmonics created by the rectangular switching waveform, and appears in the 4 kHz bandwidth of the radio noise meter as a cw signal. In addition to this signal, there is an impulsive signal at a level of 50 dB(uV/MHz). Since the radio noise meter impulse bandwidth (6.31kHz) is a relatively large fraction of the tuned frequency (15 kHz), the narrowband signal will appear to have a very wide response envelope, and a considerable signal level will be present at the start of the frequency range (10 kHz). In addition, less than four bandwidths away resides the high level second harmonic of the switching power supply frequency. Contrasting the above narrowband signal is the 50 dE(uV/MHz) impulsive signal. Due tothe fact that at 15 kHz we still have a 6,31 kHz impulse bandwidth, now relatively narrow compared to the reference bandwidth of 1 MHz, we can see only a very small component of this impulsive signal. The actual voltage level will be approximately 6 dH(pV), as shown by equation (1). The change in impulsive voltage level due to bandwidth difference from the 1 MHz impulse reference bandwidth can be calculated as follows: actual BW in MHz ALE = 20 log ( (1) 1MHz AdB = 20 log(O.O0631/1) A~B = -44 do Vi = 50- 44 = 6 dB().lV) One can then change both levels to voltage, add them algebraically, then reconvert their sum back to a level in dH(uV), to find the difference in meter indication that is caused by the presence of both signals simultaneously. These calculations are shown in equations (2.1) and (2.2): x dB(PV) = 20 (2.1) and manipulating to equation (2.2) y )lv= log-l(~) (2.2) First, considering the narrowband signal of 60 dB(uV) using eq. (2.2), y = log-1(60/20), we find 60 dB(HV) = 1000 uV. Next let us consider the 50 dB(uV/MHz) impulsive signal level, which we have already calculated to be ~6 dB(UV) in a 6.31 kHz impulse bandwidth, Using equation (2.2) again, y I.~V = log-1[6 dH(uV)/20], we find this level is =2 I-IV. We now can algebraically add the two volttage levels for a combined signal level of 1002 uV. The next step will be to convert back to a decibel scale to find the meter reading of the combined signals. Using equation (2.1), x dB(uV) = 20 log(1002), we find a level of ~60.017 dB(uV), showing that the 50 dB(uV/MHz) impulsive signal adds a meagre 0.017 dB to the level measured with the narrowband signalonly. d%iV) 605040302010O-lO-2oI FREQUENCY IN KHz Fig.1: Measurement in Wideband with Detector in Peak Position Herein lies a large part of our problem. With the accuracy and precision of most radio noise meters being such that a difference of 0.017 dB is insignificant, and probably unmeasurable, how do we measure a not so insignificant level of 50 dB(uV/MHz)? Let us first consider the options we would have in performing these measurements manually, then we will try to develop an automated method. The major problem interfering with our ability to arrive at a correct impulse level measurement is the presence of a narrowband signal and its associated harmonics. An obvious solution, therefore, would be to eliminate the presence of the narrowband signal, with the use of a sharply tunable notch filter, thus removing the narrowband signal from the spectrum viewed by the EMI analyzer. In the same fashion, using additional filters, one can remove the associated harmonics. Operating one frequency range at a time, being careful not to take measurements on the "skirt" of the filter characteristic, the operator could obtain valid readings on the impulse level present. This method, however, will be cumbersome and may not yield valid results if the encountered narrowband signals are spaced too closely together in frequency. Generally, however, this method can be used to arrive at reliable, valid results, If the operator does not have access to a series of tunable notch rejection filters, another method must be attempted,,The first step would be to decrease the I.F. bandwidth to decrease the frequency range masked by the narrowband signal on the skirts of the radio noise meter selectivity characteristic. The operator must note, however, that by changing the width of the I.F. bandwidth, he is sacrificing some of the impulse sensitivity of the 1 Ad -5- Receiver: NARROWBAND, CARRIER Recorded Level: -10 dB(1_lV) radio noise meter as is shown in Table I. Sensitivity Impulse B.W. 34 dB(HV/MHZ) 6.31 kHz 1.26 kHz 38 dB(FIv/MHz) 97 Hz 50 dB(lJV/MHz) Table I. Typical Radio Noise Meter Specifications Frequency 15 kHz 15 kHz 15 kHz Changing to a 1.26 kHz impulse bandwidth sacrifices approximately 4 dB of impulse sensitivity, but changing to a 97 Hz impulse bandwidth sacrifices 16 dB of sensitivity -obviously too much. From this observation we see that we cannot decrease the impulse bandwidth to less than 1 kHz and still get reasonable impulsive sensitivity. The operator must note, however, that when using a narrower bandwidth, he must apply the appropriately increased bandwidth correction factor to reference to a 1 MHz bandwidth. The bandwidth correction factor calculation is shown in equation (3). where x = correction factor and y = bandwidth used (in MHz) x = 20 log(1 MHz/y) (3) Now that the operator has narrowed the frequency range affected by the skirt of the narrowband signal, he can tune to a point where the narrowband level is a significantly lower portion of the total signal level measured. (The operator must note that the impulse level will have also dropped, probably by about 14 dB, but the narrowband signal level Will have generally dropped significantly more.) The next step that must be performed by the operator is to identify the impulse and narrowband portions of the signal. To do this the operator should change the radio noise meter detector function to a carrier or average detector, tune the receiver to the lowest possible amplitude point on the narrowband signal, and take an amplitude reading in d.B(HV). dB&') FREQUENCY IN KHz Fig.2: Measurement in Narrowband with Detector in Carrier Position The operator must then change the detector function back to peak and take another reading in dB(pV) (See Figure 3). The readings can then be converted back to voltage levels, algebraically subtracted, and reconverted into dB(pV) and dB(pV/MHz) levels respectively. An example of these calculations is as follows: Receiver: NARROWBAND, PEAK Recorded Level: -3 aB(uv) where x = level in lJsr and y = level in dB(HV) x = log'l(y/ZO) For Peak reading converting to HV: x = log'l(-3/20) x = 0.7079 yv For Carrier reading converting to HV: x = log'l(-lO/ZO) x = 0.3162 HV The difference is 0.7079 HV - 0.3162 I_IV = 0.3917 nV. Converting the difference to dB(HV): y = 20 log(o.3917/1) y = -8.1 dB(HV) Calculating bandwidth correction factor using equation (3), x = 20 log(1/0.00126) x = 58 dB Impulse level in dB(yV/MHz) = level in dB(l.lV) + bandwidth correction factor, or -8.1 + 58 y 50 dB(HV/MHZ). dG!J) 6050403020toO-IO- I -2o-, IO I :5 nw&i3 I I I I IN25~H~ :o :i Fig.3: Measurement in Narrowband with Detector in Peak Position Now that we have discovered a viable method of performing these measurements in a manual mode, we must explore a means to arrive at the same results using an automated system, Through experimentation it has been found that the second manual method lends itself very well to automation. The automated system can identify the frequencies of the narrowband signals by monitoring the FM video output. The system can tune through a frequency segment, identify, locate and measure all narrowband signals, storing the data as it goes. Once all signals areidentified in a particular segment, the computer accesses the data it has stored, and analyzes the signal pattern it has encountered. During this analysis the computer locates the best possible frequencies to attempt to gather valid broadband readings. Once this procedure has been completed, the computer then tunes the receiver to the first selected position, and a reading is attempted. A narrow bandwidth is selected, and data is taken first with the detector in the peak function, and then with the carrier function. When data acquisition is completed at this frequency, the computer checks the collected data for a measurable difference in the peak and carrier levels to determine if it is feasible to arrive at an impulsive signal amplitude. Should the analysis show that it is indeed possible to arrive at a -6- valid reading, the computer calculates the impulsive signal level in the manner previously described, stores the calculated data,proceeds to the next previously selected frequency point, and repeats this procedure until all such points are completed in the frequency segment. A problem arises, however, when the computer analysis determines that a valid impulse level cannot be arrived at by the previously described algorithm, At this point the computer notifies the operator that it cannotproceed with calculations at this frequency point and that further manual investigation is necessary, Once the message has been noted, the computer proceeds to the next point to be analyzed. After all data collection and analysis have been completed, the computer adds tranducer factors and other correction or calibration factors, if any, and plots the data against the desired specification limit or reference. u r’ z References [l] Heirman, D.N., "Automated Immunity Measurements," EMC Syqosium & Exhibition, &ich, March 5-7, 1985, Session A [2] CISPR Publication 16 (1977) and Amendment 1 (1980), "C.I.S.P.R. Specification for Radio Interference Measuring Apparatus and Measurement Methods" [3] ANSI C63.2 (1980), "American National Standard Specifications for Electromagnetic Noise and Field Strength Instrumentation, 10 kHz to 1 GHz" [4] NIL-STD-461B (1980), "Military Standard, Electromagnetic Emission and Susceptibility Requirements for the Control of Electromagnetic Interference" [5]MIL-STD-462 (1980), "Military Standard Electromagnetic Interference Characteristics, Measurement of" [6] MIL-STD-462-Application Note (1980), "Identification of Broadband and Narrowband Emissions," Aeronautical Systems Division, Electromagnetic andInterference Compatibility Branch, (ASD/ENAMA), Wright-Patterson AFB, Ohio 45433 [7] united Kingdom Def. std. 59/41 [8]6DE 0871/6.78, "VDE Specification, Radio Frequency Interference Suppression of Radio Frequency Equipment for Industrial, Scientific, and Medical (IsM) and Similar Purposes" [9] Rules & Regulations of the FederalCommunications Commission (FCC), Part 15, Subpart J, "Computing Devices," and FCC/OST MP-4 (1983), "FCC Methods of Measurement of Radio Noise Emissions from Computing Devices" [lo] van Essen, J.C., "Instrumentation of Automated Electromagnetic-Compatibility TestFacility for Space-Craft," EMC Symposium & Exhibition, !&rich, March 5-7, 1985, Session A @.l]Eumurian, KG., "Computer-Aided Control of EMP Measurement on Large Scale Sys_. terns,"EMC Symposium & Exhibition, Ziirich, March 5-7, 1985, Session A . -E-z3 .-L____-_______J: --c_-_______ t: still further, but the number of operator interventions can be greatly decxeased by using this automated procedure. The technique described above is in opposition to that suggested by the ASD application note ES], but it provides the correct data under non-ideal measurement conditions. The approach suggested by the application note cannot provide the correct data under similar non-ideal measurement conditions. Also, we have not addressed the proper application of transducers, so if the mandatory measurement method requires a theoretically unsound use of a transducer such as an antenna, we may still be collecting much incorrect data. : . _-__-_______ ----_I-_____ -30 IO 20 FREQUENCY 30 IN 30 20 IO FREQUENCY IN 35 KHz 35 KHz Fig.4: Automated Systemp;;;dband/Narrowband Proceeding in this fashion, the computer (controlled radio noise meter) can perform a complete EMI emissions test, collecting large quantities of data and undertaking an immense number of mathematical calculations, in a relatively short period of time. Unfortunately, there will be situations encountered that require the judgment of the human operator, showing that inquiry and development must proceed 2A2 -l- AUTOMATED IMMUNITY MEASUREMENTS Donald N. Heirman AT&T Information Systems Holmdel, New Jersey 07733 USA Automation of EMC Testing should be viewed as an engineering tool and not as a replacement for the engineer who must determine compliance with either regulatory or corporate EMC criteria. As a tool, automation if implemented properly decreases the likelihood of measurement error due to operator inattention, test instrumentation misadjustments, and inability to recreate all the test conditions on a repeatable basis. The cost of automation must also be weighed against the increased test time normally associated with manual operations. This paper will address the proper use of automation in immunity testing. The areas where automation is most useful are shown by describing a typical immunity test using a transverse electromagnetic (TEM) cell. Introduction In recent years, the proliferation of a wide range of RF noise sources from commercial broadcast stations to microprocessor-controlled appliances have increased concern for product susceptibility. Of course, in military systems, the need for product immunity (the positive view of susceptibility) is vital for strategic and tactical systems. On the other hand, consumer product immunity is generally designed to respond to pressures of the market place. A too sensitive product to the RF ambient would cause customer complaints and lead them to purchase a competitor’s product. The sheer magnitude of immunity testing has created much automation in testing in an attempt to meet production schedules and to ensure that the product has been made immune to all sorts of RF environments. The advantages and in some case the disadvantages of automation of susceptibility tests are presented from the viewpoint of the test engineer involved with consumer products, Automating Engineering Evaluation Stage Even before a product is well along towards prototype or preproduction models, testing can be used to assess the relative immunity of the product during the development cycle. At this time, it is more important to get sufficient data in a short period of time to evaluate immunity progress. This phase is generally called engineering evaluation. Here automation can provide a quick view of product immunity. All the test parameters can be held constant from test-to-test, especially when instrumentation is computer-controlled and the product response automatica(\y recorded. During engineering evaluation with incomplete or laboratory models, it is generally more important to see if there is any immunity response at all with the minimum test time. Typically, levels higher than the anticipated RF ambient are applied with a frequency scan rate faster than that for a full response of the equipment under test (EUT). The higher field, faster scan is traded for a lower (and perhaps closer to the design immunity limit) level and slower scan for full response. Here automation is a requirement since an operator may not be able to keep up with all the necessary instrumentation settings and EUT monitoring. Response algorithms based on scan rate and frequency response can be written to guide the chosen scan speed. These algorithms can be used and evaluated to ensure that the final compliance test is truly respresentative of the product immunity. The engineer evaluation period also provides a time to fabricate sensing hardware and to adapt automation software to determine better the actual EUT performance degradation as a function of applied level, scan rate, and type of applied signal (AM, PM, FM, impulse, etc.). The need for automating the remote operation of the EUT is also studied during this time. Such operation might be controlled by remote computer terminals, load simulators, or the instrumentation controller itself. If mechanical operations are needed, pneumatics may be used. Recreation of the Immunity Field The RF environment is a complex one in both time and frequency domain. Electronic products generally respond undesirably to certain frequencies and waveshapes, not the aggregate. This response is documented primarily by studying interference cases or by testing to several representative ambient signals. Unless specifically designed for immunity, commercial electronic product performance can be expected to degrade at some point during the life of the product. The seriousness of the degradation may or may not warrant design or in-the-field changes. Examples of degradation include: 1. 2. 3. 4. 5. 6. 7. Increased bit error rate Erratic Operation Abnormal modes of Operation Audio Rectification Seize up modes M,iscalibration Component failure There is always a problem with the ability of any transducer to recreate the in-use praduct RF envlranment. The literature in the USA shows that the vast majority of RF data taken is associated with commercial broadcasters and other licensed radio services.[l,2,3,4]. Hence most immunity tests attempt to recreate these narrowband radiated fields. There &much less data on RF conductedinto the product via the AC power mains and other signal or interconnecting cabling.[5] There is even less data from impulsive or aperiodic signals produced by switching transients and other localized fields such as that from a cooling fan for a power supply. Of course, special RF surveys can be made to better describe the actual ambient at product locations. This requires considerable time and must be funded for proper instrumentation. As a conse- quence, mOSt manufacturers rely on data already contained in the literature in setting up immunity test levels. System Immunity Test A typical immunity test program would include both radiated and conducted tests to include the following: Radiated 1. 2. 3. 4. Electric Field Magnetic Field Electromagnetic (Plane Wave) Field Impulsive Noise Conducted 1. 2. Immunity Immunity Direct Coupled Near-field coupled The radiated test would expose the entire product to a radiated field. The associated peripherals, l/O cables and other subsystems would also be simultaneously exposed. The conducted test concentrates on powerline and signallcommunications lines. The above list is not exhaustive. It is clear that any automation to help relieve such an extensive test program activity is highly desirable. To focus our attention on one of the most used tests, this paper will concentrate on automation of radiated immunity tests where the radiated filed is a narrowband electric field typical of AM, FM, or TV broadcast fields. The basic instrumentation for creating broadcast fields in a controlled chamber is comprised of an RF generator, modulation source, power amplifier, and transducer (antenna). Since broadcast transmitting antennas are generally far enough removed so that a plane wave is incident on the product, the presence of the product does not cause the transmitting power to increase or decrease. However, the field in close proximity to the product, does differ from that with the product removed. Most RF environment surveys measure the field with antennas removed from any object that would affect the measurement including the ground, i.e., the antenna is located several wavelengths above ground except for AM broadcast. This is close to measuring a free space value of field strength. Hence, we want to recreate that fiefd in a controlled manner. All such RF environment simulations have the potential for errors. Immunity measurements even at open area test sites must contend with and account for the ground reflection. Measurements made in enclosed chambers have even more reflections if the walls are not anechoically treated. That leaves few choices of test facilities that readily approximated free space. One choice is a parallel plate capacitor (stripline antenna or a transverse electromagnetic (TEM cell). Both provide a plane wave for frequencies within the passband of the transducer. RF anechoically treated shielded rooms (all six surfaces) are yet another choice for free space measurements. However, anechoic chambers are usually more expensive. Oncethe type of measurement facility is selected, a method for automating the recreation and the monitoring of the immunity field can be implemented. Generation of the necessary fields are relatively straightforward and will not be discussed further. The monitoring of these fields is not straightforward and great care must be exercised in monitoring the field around the EUT. The most popular monitoring procedures are: 1. 2. 3. Real time leveling using a field probe next to the EUT. Recalling from controller memory signal source drive power based on previous measurements of field strength in the test volume with the EUT removed. Recalling from memory the source drive to set a desired field strength based on the calculated field using antenna gain, radiation pattern, and signal level input. Item 1 has the potential for monitoring a field that is largely affected by the EUT, especially at frequencies where the EUT resonates. Items 2 and 3 are preferred if the EUT does not significantly interact with the transducer to affect the calibration of the applied signal. Both of these latter items to be fully implemented require automation to look up the calibration data and control the signal input into the power amplifier. A Sample Automated Susceptibility Test To further focus on the benefits of automated immunity testing, we describe a typical test using a TEM cell as the radiating transducer for launching an RF narrowband electric field ambient. Figure 1 shows typical instrumentation. TEM cell testing provides a passband of operation from dc to a frequency where the dimensions of the cell are approximately equivalent to a wavelength. For a cell with dimensions as shown, the useful upper frequency is about 165 MHz for EUTs with dimensions of up to 10 by 30 by 30 cm. General test guidelines are contained in Reference 161.We now expand those guides for this example. First, the EUT dimensions should be kept small compared to the dimensions of the cell’s test volume. If not, errors in the applied test field will increase due to the capacitive loading of the EUT. Generally the linear dimensions of the EUT should be kept to no more than about 30 percent of the associated test volume dimensions in either the top or bottom half of the cell. The far field immunity level at the center of the test volume (midway between the center conductor and ground plane) can be calibrated by several methods with the EUT removed: Monitoring input RF voltage using a monitoring Tee for frequencies typically below AM broadcast frequencies. Monitoring net power flow into cell using incident and reflected power and a bidirectional coupler. This technique can be used for all frequencies within the passband of the cell. Monitoring the electric field directly with probes. The first two aproaches are accomplished external to the cell which has distinct advantages since no cables exposed to the high RF field. Automation is virtually a necessity to keep track of these levels and to perform net power flow calculations as well as repetitively calibrate the meters. The last approach requires the most care. In the probe approach an optic link is generally required to not disturb the field or become a radiating or scattering structure. The placement of the probe is also critical since at EUT resonance, for example, the field is most perturbed and a probe in the near or scattered field will indicate fields that are different from the nomimal test level. During the actual immunity test, the fields can be monitored using the above three basic approaches. The levels will differ from nominal due to the loading affect of the EUT. If the dimensions of the EUT are kept to the 30 percent test volume criteria, the level differences from nominal will be in the order of &36 dB under cell multimode frequency. The most useful way to evaluate what is happening to the electric field is by using an electrically short dipole or monopole probe. To avoid the near field scatter problem at EUT resonance, these probes can be placed in the half of the cell not occupied by the EUT and at a point which is the mirror image (about the cell’s center conduct) of the geometrical center of the EUT. Above multimode, placement of the probe becomes much more critical to remotely monitor the field at the EUT. The differences between the nominal immunity level and what is read by the probe significantly increases making this monitoring method less useful. Characterizing the effects of all the monitoring methods is a useful undertaking. For example, one of the benefits of such probing may be to extend the useful upper frequency limit of the cell or to allow larger EUTs. In practice, the TEM cell can 2A2 BIDIRECTIONAL MODULATOR CONTROLLER - APPLIED - TELEMETRY/CONTROL IMMUNITY SIGNAL LEADS LEADS PERFORMANCE DEGRADATION SAMPLE NOTE: ElJTlPROBE MONITORING CABLING VIA TEM CELL FIBER OPTIC OR HIGH IMPEDANCE LINES INSIDE CELL. Figure 1. Typical TEM Cell Immunity be used above its normal upper frequency limit. In this frequency region, the field strength is complicated by the multimoding of the cell. Only through use of automated data gathering techniques can the cell be properly mapped to determine the field throughout the test volume. The mapping would be much too cumbersome using manual techniques for recording the orthogonal (and total) field components. In this case automation is the only practical way to extend the test capabilities of the cell. Performance Degradation Monitoring The next area where automation helps is in recording performance degradation as a function of applied field strength, frequency, modulation, degradation type, EUT response time, etc. Much of this is simple data bookkeeping. However, there still persists those who want to visually determine performance degradation. If degradation monitoring were constantly done by this means, especially by viewing a CRT, errors will soon occur due to the long, repetitious and boring nature of immunity tests. No matter how conscientious the operator, monitoring of anticipated, slow to materialize, visual EUT degradation is prone to errors and lack of repeatability. Typical automation of performance degradation would include monitoring analog signals directly onto the IEEE 488 general purpose bus or digital information on an RS 2326 interfaces DIMENSIONS I = 200 cm w = 95 cm h = 65 cm Instrumentation cable. These methods: signals are routed to the bus by one of 2 a. Direct connection to monitoring points within the EUT via fiber optic or high impedance transmission lines. b. Indirectly via connection to EUT performance monitors, external controllers, external circuitry, simulated loads, or peripherals, all of which are not in the test chamber but are connected to the EUT via cabling. The former method requires several telemetry links not part of the EUT. The latter relies solely on using part of the EUT system that is not exposed to the high fields, except of course for the interface cabling inside the test chamber. Proper filtering of these leads through the TEM cell walls are needed to protect the equipment outside the cell from RF on the cables extending through the walls of the cell. Once the degradation is recognized by the computer, preprogrammed operations can be implemented. Some operations are shutting down the amplifier if a destructive level of degradation is reached, sequencing to other EUT modes of operation, and pausing on particular frequencies to evaluated EUT response time to the applied field. Other instrumentation activities can also be conducted while immunity is being recorded. For example, in the TEM cell, 10 - there are relatively high cell Q’s above its normal upper frequency (multimode) limit, the output of the signal source power amplifier chain should be filtered so that the second and higher harmonics are suppressed by at least 60 dB. This will avoid a false EUT response at the signal source frequency when in fact the response is due to a harmonic of the applied signal (generated by the amplifier) which is coincident with a multimode response. Automatic switching of low pass filters is a necessity since the test engineer is concentrating his attention on the EUT degradation and operation and could easily forget this switching detail. It cannot be overemphasized the importance of spending the extra time to automate the performance degradation monitoring. The test controller can do most of this, especially if all degradation can be sent to the controller using analog (via an A/D converter) or digital (via the IEEE 488 bus or EIA RS-232 telemetry) signals. The test engineer should where at all possible take advantage of performance monitoring by sensing signals on the same leads which remotely operate or communicate with the EUT from outside the test chamber. This will avoid introducing additional cabling which itself might be vulnerable to the applied field causing a false degradation indication. Final Immunity Evaluation Once engineering evaluation of the product immunity is finished and suitable mitigation applied, the final compliance test is performed. This test must be highly repeatable and calibrated to judge compliance. Here automation will significantly increase the test repeatability and ensure that separately derived calibrations are always used. These tests are generally longer in duration since the full range of performance degradation is checked and recorded for the final test report. This phase is particularly methodical and a great deal of degradation bookkeeping is necessary. For example, the frequency scan rate may be varied to ensure that a degradation response is not missed. The affect of the complex field within a TEM cell above multimode has to be accounted for here if used. It may be necessary in the multimode range to move the EUT in the cell to expose various circuitries to the full field gradient caused by the standing wave pattern which can amount to field uniformity errors in the order of 10 dB or more. Even under multimode, there are undesirable TE and TM modes launched that at the very least should be accounted for in the measurement error. All of these factors are best recorded and controlled via a well thought out and planned automation program. Conclusions This paper described the usefulness and precautious of automation of immunity testing. Automation if used properly is a powerful tool that can be used to produce a test with less operational errors. However, automation which is not periodically checked by manually performing a test, tends to lull users into a sense that the results of such tests are irrefutable. Periodically it pays to manually set all instruments and see if the results are the same as that found by automation. The paper has also shown the concern for ensuring that the EMC engineer correctly automates the immunity test to replicate the appropriate immunity field and to monitor the proper perform degradation. References PI D. E. Janes, R. A. Tell, T. W. Athey, and N. N. Hankin, “Nonionizing Radiation Exposure in Urban Areas of the United States,” Proceedings, IVth International Radiation Protection Association, Vol. 2, pp 329 - 332, April 1977. 121 R. A. Tell and N. N. Hankin, “Measurement of Radiofrequency Field Intensity in Buildings with Close Proximity to Broadcast Stations,” U. S. Environmental Protection Agency Report ORPIEAD-78-3, August 1978. 131 D. N. Heirman, “Broadcast Electromagnetic Interference Environment Near Telephone Equipment,” IEEE National Telecommunications Conference Record, Catalogue No. 76 CH 1149 - CSCB. 141 G. Costache et al., “Electromagnetic Field Strength Probability Profiles for Canadian Cities,” International Electrical and Electronic Conference and Exposition, Toronto, Canada, October 1981. [51 FDA Medical Device Standard, “Electromagnetic Compatibility Standard for Medical Devices.” MDS-201-0004, October 1, 1979. PI M. L. Crawford and J. L. Workman, “Using a TEM Cell for EMC Measurements of Electronic Equipment,” U. S. National Bureau of Standards Technical Note 1013, April 1979. 3A3 1. Abstract The purpose of this paper iS t0 give a complete overview of an Automated EMC Test Facility in operation, for Emission-, Susceptibilityand Time domain measurements. The contents include system set-up, specifications and drawings with a description of different test set-ups used for spacecraft, subassembly or unittesting. The narrowband and broadband aspects are highlighted, and a plot of the data output from the system are included. Conclusions are drawn with respect of specifica' tions, test time, accuracy etc.. 2. History Since the time when the basic idea of automating the EMC Test Facility was conceived and initial funding became available, the line of thinking had changed quit a bit. Due to improvement of the test equipment, measuring techniques and the budgetary constraints, the original idea of setting up a separate system for Emission- and Susceptibility- Testing had to be abandoned. Instead, a new design was set up in such a way that all instrumentation performs a multiple function and will be used for Radiated/ Conducted Emission and Radiated/Conducted Susceptibility. The existing test equipment is integrated in this system as well. A block diagram,, Fig. 1, shows the basic set up. Due to the fact that a broadband high power requirement will increase the cost of the radiated susceptibility part of the system by 100% or more, it has been decided to keep the requirements of 30 - 60 V/m and to accommodate projects with the necessary power at a given frequency "narrowbanded", which complies with the experience so far. All specifications for this system are derived from spacecraft requirements existing today and in the near future. The system has been designed to meet these requirements. 3. Introduction The system will be used for the following measurements: A) Radiated Emission Measurements over a frequency range from 20 Hz - 40 GHz, B) Conducted Emission Measurements over the frequency range from 20 Hz - 100 MHz, C) Radiated Susceptibility Measurements over the frequency range from 20 Hz - 40 GHz, D) Conducted Susceptibility Measurements over the frequency range from 20 Hz- 100 MHz, With the possibility of injecting CW and pulsed signals, to test power-, signal- and commande lines. The block diagram of Fig. 2 outlines the complete system set up. FIG.1 x- TEST -FACILITYGENERAL SET-UP All equipment used in this system is operated to IEEE-488 standards or equivalent. The possibility of opto-coupler extention is provided for operation in a radiation-hazardous area. The System Controller is a desktop model with a large screen, so the facility engineer is able to program it and to modify the software during the test. For the sweep section we stay as long as possible co-axial, in order to facilitate the test work. However, a synthesizer is a must, due to the frequency accuracy required. The exact specifications of each subsystem will be discussed separately. For the Radiated Susceptibility testing we have for the Low frequency range, Amplitudemodulation, for the Megahertz range Frequency-modulation and for the Gigahertz range Pulse-modulation. - 2 PRINTER /_ HP-9876 A + P-F9 HP-B112 A w CURRENT-FKE?E SOLAR. 6741 1 0: PULSE GEN. MOO SOURCE I $u-6. N &JNO 4. SCKUM AN4LmR HP-B566A lCQi-4OGHz TWT 1077 H12 17 FIELD MONflOR CONOUCi% D -SUSCEPTIBILITY SETUP FOR RADIATED -I CONDUCTED EMISSION NOTE C.5W - COAXIM RL - REIAY t pp&LJ-:_a\,; L. v+ r---- --7 HP-37203 A I :_______--,‘,‘_______I I 15 EXTENDER ,’ ’ .’ ,’ RADIATED-! CONDUCTED SUSCEPTIBILITY SWITCH ESTEC %3?2iii;,~____; / \,--.’ EMC TEST FACILITY FIG. 2 _ - Amplifiers from 20 Hz - 18 GIIzwere existing in the EMC Test Facility and are integrated into the system. This together with two new amplifiers; one from 18 GHz - 26.5 GHz - "K" band and one from 26.5 GMz - 40 GHz - "KA" band. In this way only one waveguide and standard gain horn will have to be used for each amplifier. All cower for the radiated susceptibility test will be fed through a dual directional coupler to the antennas. In order to control the levels on the antenna a Dual-Sensor-Powermeter is used in order to measure set-power and reflected power. For the radiated- and conducted emission a front-end receiver is used. It has to supply the necessary pre-amplification over a frequency range from a minimum of 100 Hz to 18 GHz, and preselection below 2.4 GHz. The unit will be used with the HP-8566 spectrum analyzer. Blockdiagram Fig. 3. This pre-selection below 2.4 GHz increases vulnerability to overload, especially on broadbandnoise, and maximizes sensitivity, while at the same time maintaining the highest possible instantaneous system bandwidth and dynamic range. In addition to this receiving system a broadband electric field antenna is used, working over a frequency range from 20 Hz - 1 GHz. Apart from the existing spectrum analyzer, a second analyzer has been introduced, partly used for susceptibility testing. This in order to control the injected voltage and current on the line under test. Due to the fact that scientific spacecraft are low-noise, but still produce noise with its own frequency spectrum, the susceptibility levels to be injected are small, depending on the applicable voltage- or currentlimit, which can be in the order of 20 dBuR. In order to link the system together several co-axial switches and relays are used, positions and number are shown in the blockdiagram of Fig. 2. To control system operations and monitor the behaviour of the U.U.T. (unit under test) a digital voltmeter combined with a data acquisition unit is used. The data acquisition unit is important for the system, it includes: channel multiplexers, relay multiplexer, voltage- and current DAconverters and a real time clock. Also, time domain measurements are an integrated part of the test activities. A bus controlled oscilloscope is used for this purpose. And, last but not least, a printer iS intergrated with a dump feature, which gives out the test data, like: a narrowband plot containing the narrowband signals only includiny spec-level and frequency printout with measured levels. An example of a plot is included. The same can be achieved for broadband measurements or conducted- and radiated susceptibility measurements. In fact this system produces a complete testreport with detailed information directly after completion of the test. 4. Description and Specifications In this section the system will be described with reference to blockdiagram Fig. 2, and broken down in subsections in order to have a better overview of the system. 13 3A3 - IF-DISPLAY SECTION RF SECTION I Ym.l.b.___._L__=i._ .I7 I_.__._____-__.-.___._.d.q d1.4 :m-? 521.4 MHz IF IN MHz IF OUT :...._..._.......__..____....................................: FIQURE 3 MOOEL 254EC RECEIVER SYSTEM. BLOCK DIAGRAM 4.1 Control part The "control part" consists of a HP-9836-S calculator and a HP-9876-A printer. The main reason for using a "desktop" calculator is that the controller can be handled and programmed by the facility engineer directly, without having to ask for software support, which means no loss of operating time, maintaining and updating of a more complex system Apart from the above, it:is of utmost importance that the "EMC-Engineer" on the job can translate his EMC problems directly into the machine, without external software support. This in order to avoid unnecessarily complications of already complex problems. The calculator has a 12" CRT and two built-in disc-drive units for 5%" floppy disc. Memory capability expendable up to 2 Mbytes, with Basic,(extentions) Pascal, graphics dump and storage. This system is also fitted with an extra HP-IB and BCD interface. More than 4 bus expanders can be added to provide 16 additional slots for memory- and I/O cards. In addition there is an HP-IB extender with fibre-optic interface. The use of fibre-optic links has a special meaning for this system. Due to the fact that in our case the system isa combined one, and used as well for "Emission Measurements!' as "Susceptibility Measurements". It is very important to separate the transmitting and receiving sections to avoid unnecessarily problems due to small beaks, cable- or small ground loops, which would limit the dynamic range of the equipment. The use of fibre-optic links has imporved EMC measurements quality considerably and is now standard in our facility. The thermal graphics printer can handle a graphics dump from the 9836-s CRT within 10 seconds. In this way, the test data are available seconds after the test, a protection against nailbiting and nose eating customers who are nervously waiting for the test result to be produced, to see if they are in- or out of specifications. 4.2 Receiving part The receiving part consists of two units: A- SRD-2548-C precision wideband front-end receiver. B- HP-8566 spectrum analyzer. A more detailed blockdiagram is given in Fig. 3. The specification is the combination of the two instruments. Up to 2.5 GHz pre-selection is obtained by pre-amplification. Above 2.5 GHz we have Yig-pre-selection. Use of this front-end receiver in combination with the spectrum analyzer has imporved the sensitivity, dynamic range and instantaneous band width of the spectrum analyzer without losing any of its features. The system has been set up to operate over the frequencyrange from 20 Hz - 40 GHz. However, from 20 Hz to 100 Hz, extra care has to be taken due to the fact that we have to work so close to the local oscillator and having to extrapolate the antenna factor. The combination can perform measurements up to 18 GHz with a typical noise figure of 10 dB and a dynamic range from 72 dE to 1 MHz BW. From 18 - 26.5 GHz the analyzer is used with a harmonic mixer type HP11970-K and from 26.5 - 40 GHz with a harmonic mixer type HP-11970-A. With a noise level of approximately -110 dBm by 1 KHz. Bw. For equipment layout see Fig. 4. In addition to Fig. 4 we have Fig. 5. Showing the same set up, but with the HP-11517-A Bias Mixer. This has the disadvantage of 20 dB less sensitivity, plus the fact that each frequency line has to be investigated. Must be manually adjusted. Additional software-driven routines allow for automatic calibration of external sensors, such as antennas and current probes. The calibration routines will accept and store externally derived calibration data. Another important feature included is the overload sensing and warning in all the critical areas of the RF signal path. If a signal overload condition ever occurs, automatically the signal gain will be decreased in the appropriate part of the system. In the normal remot digital control mode the interface connects directly the receiver with the analyzer. In this way the receiver controls the analyzer and the entire receiving system need only appear as one device for purposes of addressing and control. In this role the system is both a listener and a talker. All data transfer functions from the spectrum analyzer display section are retained. Fig. 5 The combination of the calculator, front-end receiver, and spectrum analyzer is the most powerful tool for EMC measurements I have seen sofar. It is able to step from one frequency line to the next and evaluate each data point for narrowband or broadband criteria (according to Mil-STD in our case). If necessary at the same time coherent and incoherent broadband noise can be separated, and narrowband and broadband data can be graphed on separate plots and each individual frequency point can be printed out. For analyzing the test results is this a very important piece of information. In our set up a narrowband emission plot will take about 30 minutes (20 Ilz - 1 GHz) Fig. 6 This in combination with a relatively new type of receiving antenna, type SAS-1D from Antenna Research Associated Inc. (Fig. 6) It is an electric field antenna over the frequency range from 300 Hz - 1 GBz. The low-band circuitry is such that the response rolls off at the rate of approximately 20 dB per decade of frequency below 300 Hz. The system consists of two electrically separate antennas, namely a top-loaded monopole for low-band and a discone for high-band. The yolarisation is vertical and the directivity - 15 is omnidirectional. Overload for 1 dB comperssion: Lo-hand 0.5 V/m, Hi-band, 0.1 V/m. The antenna is pOrtable and especially suited for indoor applications. For all our conductive measurements we use the well known Solar current probe type 6741-1, frequency range 20 Hz - 100 MHz, which has a flat frequency response over the frequency range 10 KHz - 100 MHz. Maximum current: 300 Amperes ac or dc;load Impedance: 50 + j 0. ohms. Direct connection to the conductor is not necessary, since the probe may be opened for insertion of the conductor. 4.3 Sweep Section To cover the frequency range of 20 Hz-40 GHz required for this system, three instruments are used: A- HP--8165-A Frequency Synthesizer and Function Generator. 0.1 Hz-50 MHz. B- BP-8673-D Frequency Synthesizer. 50 MHz - 26.5 GBz. C- WJ-1204-40 Milli!meter-Wave Frequency Extender. 26.5 - 40 GHz. The HP-8165-A programmable signal source is a versatile function generator with good accuracy. Microprocessor control ensures rapid programming amplitude output from 10 mVpp 10 VPP, amplitude- and frequencymodulation. The HP-8673-D synthesized signalgenerator has precise signal simulation capability. The frequencies are derived from a quartz crystal time base, via a direct synthesis technique providing extremely low signal sideband phase noise. Harmonically related spurious C-60 dBc.SSB Phase noise<-80 dBc. 10 KHz offset +6 dBm output level at 26.5 GHz. at 10 GHz Leveled calibrated output to -100 dBm. Amplitude, Depth 0 - 908, pulse on/off ratio: >HO dB and frequency modulation maximum c peak deviation is smaller than 10 MHz or (see data sheet). All functions are programmable, including frequency output and RF level setting (in 0.1 dB steps). The same synthesizer is used to feed the Watkins-Johnson frequency extender WJ-1204-40. The most notable feature of this system is its excellent frequency resolution, accuracy and stability. Input power 0 dBm, output power +3 dBm. As stated in the introduction, one of the aims was to stay as long as possible co-axial, in order to facilitate the test work. However, one has to realize that starting from 12 or 15 GHz and going up, the attenuation is increasing tremendously and special attention has to be payed with respect to the length of cable, connectors etc. For this and several other reasons we kept our equipment as mobile as possible and derived a great benefit from it sofar. 4.4 Modulation part This part consists of two instruments which are used for multiple purposes, such as radiated- and conducted susceptibility testing, for testing as modulation sources and also for conducted spikes and commandline testing. From the HP-8116-A, pulse function generator all functions are bus controlled and provide sinewave, squarewave and pulses over the frequency range 100 mHz to 50 MHz, pulse width: 10 nS - 999 mS. Amplitude 10 mVpp to 16 Vpp. 3 - A3 The second instrument in this section is the HP-8112-A, programmable pulse generator with the following specification: : 950 m sec. - 20 n sec. Pulse period : 950 m sec. - 65 n sec. Pulse delay Double pulse : 950 m sec. - 20 n sec. Source impedance : 50 ohm. Output voltage: +/- 8 v into 50 ohm. 4.5 Power Meter Here we use a dual sensor power meter type HP-438A, with a frequency range from 100 KHz26.5 GHz, using the HP-8485-A Thermocouple power sensor. This has been introduced to ._/ BLOCK DIAGRAM OF THE 438A AND ITS TWO SENSORS Fig. I control the power and reflected power on the transmitting antennas. Measurement modes are A, B, A-B, B-A, A/B and B/A. The power range is sensor dependent, dynamic range 50 dB. The use of the power meter is entirely based on Mil-STD testing, which implies that the electric field is calibrated with the transmitting and receiving antenna one meter apart in an empty room, the empty room being the EMC Test Facility covered with absorbing material, in order to reduce reflection. Power levels are taken and stored in the calculator and called up during the test to set the levels. Diagram on Fig. 7. A software routine is set up to determine if the antenna is radiating, check the level and compare the reflected power etc. The been I_- use of electric field sensors has-completely abandoned, under the assumption that no source will increase the radiation power in order to satisfy the form-factor of a given object. 4.6 Monitor part This part consists of 4 separate units. Blockdiagram Ref.nr. 11, 12, 13 and 14 (Fig. 2). First instrument in sequence is the HP-8566-S, spectrum analyzer with a frequency range from 100 Hz - 22 GHz, using external harmonic mixing with a frequency range up to 40 GHz. Amplitude approximately from -137 dBm to +30 dBm, resolution 0.1 dB. Dynamic range 95 dB. Accuracy 2.2 dB over the frequency range from 100 Hz - 22 GHz. With internal software routines like Peakseards-Signaltrack, Signal identification Marker aided measurements-Max hold and saving of control setting. The above mentioned features are a must if a - 16 - ESTEC E ELECTRIC real EMC-Minded program has to be set up. Ref.nr. 12 represents the Data Acquisition unit HP-3947-A including the extender unit. 5% digit DVM which may be programmed for 300 readings per second (3% digit mode) or 50 readings per second (5% digit). It consists of a 40 channel relay multiplexer with a power rating of 1 VA per channel (170 Vp max), relay contact 1 Ohm, crosstalk -40 dB, 32 channel Mercury wetted relays are added, with 100 VA per channel (100 Vp max). Contact resistance 400...mOhm. Crosstalk -30 dB. Further we have a Dual Current D/A converter with an output from 0 to + 10 Volt. These sources can be used to provide a programmable test stimulus or to control voltage programmed devices like power supplies and VCO. And last but not least, this "Data Acquisition Unit" contains a real time clock to support all data output. With Ref.nr. 13 we have our HP-3437-A, High Speed System Voltmeter, 3% digit, voltage range 0.1 - 10 Volt with more than 5000 readings/set. in peak- or RMS value. One of the more important functions of this monitor sub-section is to control susceptibility testing. Mainly, for conducted susceptibility, we have full control on injected voltage and current and are able to plotbZ-. To be in full control means y ou have your ~__ hand on the problems. All this automated equipment -seems to be a major investment, which indeed it is, but it pays back in quality, quantity and time, highly rewarding I would say. 4.7 Amplifiers/Antennas All amplifiers are well defined on the blockdiagram Fig. 2. The measuring set up for Rad.susc. is shown in Fig. 8. We work over the frequency range from 20 HZ 40 GHz. From 20 Hz - 250 MHz : 100 Watt power. From 250 MHz - 18 GHz: 10 Watt power. 1 Watt power. From 18 GHz - 40 GHz : Electric field level is shown on Fig. 9. From the diagram in Fig. 8 we can see that the EMC Community is in great need of a much more -. effective antenna between 30 - 100 MHz. Nowever, one has to consider that this is the standard available power. But from project to project it will be investigated if an effort will be made to increase the power and field level. M C Test facilities. FIELD-STRENGTH GENERRTION Fig. 8 HP-8165 HP-8673 SYNTH /GEN -0 20Hz.-50MHz SOMHz.-26GHz HP-3497-A DATA-ACQ.UISITIOA UNIT RADIATED SUSCEPTIBILITY TEST SET-UP C _?.mnll L Gk12 ?._..I LB GHZ Fig. 9 The Blockdiagram indicates also frequency range and antennas used. For more detailed information, please refer to the data sheet. Generally speaking we can say that amplifiers, couplers and antennas are harmonized to the maximum extent possible as regards frequency range and power. 5. Remarks on Measurements 5.1 Radiated Emission I would like to highlight briefly our measurement criteria. The narrowband signals are measured in 6 scans from 20 Hz - 22 GHz, in standard spectrum-analyzer settings, with respect to "span" etc. Each band is scanned with 2 different band- HPIR SYNTHJGEN -- HP-438 -A DUAL-SENSOR POWER-METER HP-9836.. S SYSTEM CONTROLLER .FREQ.RANGE-IOKHz-40GHz 3 - 17 - A3 Conducted measurements are also carried out in the time-domain. Although it seems that this measurement is not EMC related, it is very important to be able to correlate your Frequency Domain measurement with your TimeDomain measurements. The instruments used are arbitrary; in our case we use standard equipment together with a digital storage oscilloscope: Type TEK-468. BW-ZOOMHZ, buscontrolled. The use of a digital storage oscilloscope has the advantage of good triggering possibilities and thanks to the bus you are able to dump the picture on the printer, including all information. 10 MHz Fig. 11 widths a factor 10 apart, and amplitudes from the first and second measurement are compared with a 3 dB criterion in our case. Correction factors are added before printing the signal output. Fig. 10 shows a narrowband plot. Broadband noise is measured in 4 scans from 10 KHz - 22 GHz. Scans are made in the Peakhold mode with a relatively wide bandwidth. The time set to fill each "Bit!'is equal to the data from an impulse signal with a separation of 50 Hz. Each data point contains half impulse bandwidth, and stored in a temperaly file, is compared with 6 data points before and after; if 4 3 dB (in our case) it will be processed. Fig. 11 shows a broadband plot. 5.2 Conducted Emission For conducted emission in the frequency domain we opt for the same criteria of signal processing as we did for radiated emission. Measurements are carried out on power-and synchronization lines in differential- and commonmode in voltage and current. Data- and command I lines are tested in bundles of wires, separated like; all digital input lines. all digital output lines. all serial digital data. all analogue lines. The signal ground is always tested separately, due to the importance with respect to the quality of collected- or transmitted data. Further more I would like to draw your attention to what we call a "Structure noise Test". Fig. 12. This test supplies us with important information, like leak resistance and capacitance, loops etc. This is important when making the final analyses about the unit under test. Fig. 12 CONOUCiED EM STRUCTURE -- NOISE CURRENT 5.3 Conducted Susceptibility _-.-__ -____ Conducted susceptibility testing is carried out on Power- and Sync. lines both for CW and Spike signals. Voltage and Current are controlled and checked against the limits. Voltage and Current injected are plotted together with the Impedance. If the unit under test is susceptibleted, impedance is an important piece of information to analyze the problem. Fig. I3 shows a typical output plot from a conducted emission test. GIOTTO dBuR/dBuV ESTEC EMC TEST FflCILITY CONDUCTED SUSCEPTIBILITY JPR DPERRTfoNRL MODE :EXP ON DRTE 21-09-1984 TEST ;:;; : 28 VOLT Fig. 13 Particularely signal- and commandlines are subject to conducted susceptibility testing. Due to their important function in a "spacecraft" a lot of our attention is devoted to test those lines. A special test box has been developed for this purpose. Fig. 14 shows the test set up. The command signal is fed through the box and the disturbance is injected via an opto-coupler, in order not to load the circuit. In this - 18 - .way negative pulses are injected into a "1" and positive pulses into a "0". Together, this enables us to determine the safety margin of the circuit. A typical advantage of this kind of equipment set up is the ability to supply the customer with a small box in order to control the test. This box provides him with the possibility of decreasing the signal level until the susceptibility has stopped, and to increase the signal when the susceptibility has passed. At the same time the print-out will indicate the frequency range and the threshold level of the susceptibility. 5.4 Radiated Susceptibility Maybe this part of EMC testing has benefited the most of being automated. We have seen that in the case of manual operation an operator has to control the frequency with one hand, amplitude with the other, check the modulation, overload of the amplifiers, correct T.A.F., cable losses and check the susceptibility criteria of the UUT. All this control.and check functions are now taken over by the system controller. This provides an accurate and fully corrected, measurement calibrated on the spot of each frequency step. Above all one has the ability to repeat this measurement any number of times without the slightest diviation. Also here the susceptibility control box can be introduced (as discussed under cond.susc. testing). An important aspect is that the UUT is not unnecessarily overtested. With manual operation we have seen errors up to 20 dB or more due either to human errors or mismatch. Not only do we avoid overtesting but we have also the possibility of investigating the so called "Window" effect by automatically increasing and decreasing the power level. Sofor we have not seen this "Window" effect in our facility. The fact that we are working in a shielded room is the cause of other problems like non-uniformity and antenna position this can be the subject of a calibration routine. Reflections can be controlled through the installation of reasonably sized anechoic material. But this will be only effective from a few hundred megahertz. Near field problems are even more difficult. Remember that most of the requirements call for starting at 10 KHz. Building a facility where 10 KHz is in the far field is sheer UTOPIA. And near f!ieldmodeling is a sophisticated guess. Therefore the only solution to the near field problem is to move away from Low-Frequency Radiated Susceptibility Testing. Fig. 8 shows the test set up used for radiated susceptibility testing. The use of the system interrupt box and dual sensor power meter has already been explained. 5.5 Susceptibility to E.S.D. In our facility we have extremely good use for the schaffer-NSG-430 discharge gun apart from normal applications like testing of "Spacelab" equipment which is subject to discharges from 10 m-joule. Discharges which are produced by astronauts during Spacelab missions. The same test can be made on large ground stations or computers to detect bonding faulr ts and ground loops. It cuts back expensive facility time considerably. 6. Conclusion An attempt has been made to set up a system using commercially available equipment (micro processor only). Starting on automating, one is afraid to face high cost implementations. The cost can be rather limited on the basis of "growth system". It is possible to achieve a very sophisticated system by planning the cost over a long period of time. The new system should be developed in tight cooperation with an EMC engineer, and aim for achieving flexibility and growth capability. Automatic measuring techniques has brought us many additional advantages. Like the possibility of compensating for measurements errors, direct comparison of test results and use the same results for prediction. And it protects the test-facility against poor preformance. It is not possible to go deeply into the theoretical background of the topics discussed. And a routine engineer can ask many awkward questions. However, it is hoped that successful innovating, will achieve what we call "EMC". 4A4 19 - - COMPUTER-ASSISTED CONTROL OF EMP MEASUREMENT ON MAJOR SYSTEMS Gregoire EUMURIAN THOMSON-CSF ISSY-LES-MOULINEAUX, FRANCE 1. - INTRODUCTION EMP experimentation on major systems, for reasons of cost and efficiency, require accurate prior preparation of the tests. The time available is limited and most often it is impossible to resume the tests if later analysis shows inconsistencies. It is thus necessary to have an automatic system to avoid handling errors and a control assisting system which enables refining the results and adapting the theoretical experimental program to the actual situation. It enables measuring low excitation fields (100 V/m to 1 kV/m) . The passband extends from 70 Hz to 150 MHz. The sensor takes the form of a sphere with a diameter of 106 mm. All these performances (high sensitivity, wide passband to low frequencies and small dimensions) were obtained by implementing the active sensor concept. The EMP data acquisition system consists of 5 main elements : sensors, optical transmission lines, optical link processors, digitizers and a data processing system (fig.17). Principle 2. - SENSORS To obtain a minimum volume, optimize the sensitivity, dynamic range and passband of the magnetic and electrical sensors, coil-type highimpedance sensors have been used along with built-in amplifiers and correctors and also electronically matched high-impedance electrical sensors. The total dynamic range of the sensors is extended by a sensitivity change remote-control system (0 to 70 dB 10 dB steps). Figurel8indicates the main sensors as well as their dynamic ranges (between noise and maximum saturation 1evel)passband external dimensions. The sensor takes the form of an internal sphexe (comprising the ground frame) surrounded by two half-spheres forming the positive and negative frames (Fig.)). The two antennas are connected to a differential amplifier with unit gain, via two highimpedance 40 dB attenuators. Att~ts-2M6 A,%.&lad6 Figure 1 - Principle and geometrical characteristics of free-space sensor (E30) 2.1 - EXAMPLE OF A SENSOR 2.1.1. Free space electric field sensor. This type of sensor is used to determine the incident field and the field within cavities. It has a full-scale sensitivity from 2 1 V/m to 3.16 kV/m with a total dynamic range of 120 dB (50 dB of instantaneous dynamic range and 70 dB by switching). The differential stage is followed by a 20 dB low-impedance attenuator, an amplifier (gain 46 dB)and a second low-impedance attenuator. The attenuators can be remote-controlled (via the optical link) and their sensitivity is switch-selectable over a range of 70 dB by 10 dB steps. - 20 - i.e. an effective height of : _-PARTICULARITIES OF THE SENSOR - Geometry Figure 1 indicates the geometrical characteristics of the sensor. The spherical shape . 'was selected to optimize the effective height of the antenna. Figure 2 indicates the distribution of the 'electric field over the surface of the sensor. The field, perpendicular at all points to the surface of the sensor, depends on the incident field to infinity (Ei) and its angle with the sensor axis (0). h 6IIrg EO (6) eff = ch+ce with the values indicated in figure 1. The effective height is : h 6Ilx (0,053)2 eff = = 6.68 mm (7) 3611x 10g x 70 x lo-l2 -Connection of the sensor to a high load The utilization of attenuators and high impedance differential amplifiers (Re = 22 MO) enables reducing the lower cutoff frequency to less than 100 Hz. F B - 1 2 II (Ch + Ce ) Re - = 69 Hz (8) An output voltage directly proportional to the incident field is thus obtained (non-derivative response). Figure 2 - Electric field on the surface of a metal hemisphere The value of the field is given by equation (1) : Er = 3 Ei cos g It is drawn up from two limit conditions : -zero tangential field on the surface, -field indentical to the incident field to infinity. . Figure 3 gives the equivalent diagram of each half-sensor. Ch represents the capacitance between the hemisphere and the internalsphere and Ce, Re, the capacitance and input 'resistance of the amplifier. .Figure 3 - Equivalent diagram of a half-sensor _ (free space electrical sensor) The incident field, Ei, induces a charge Q (2) within each hemisphere of radius ro : -Dynamic range For a given value of the incident field, the dynamic range depends on the effective height of the antenna as well as the noise level (N) at the input : S -= N Signal p-p = 2 Ei . heff Noise rms N (9) With a noise level of 50pVrms (100 Hz to 100 MHz band) and an effective height of 6.68mtn the dynamic range is 48.5 dB for an incident field of 1 V/m. 2,1,2. Free Space Magnetic Field Sensor The magnetic field sensors are based on a coil with one or several turns. To obtain satisfactory sensitivity to low frequencies, the section and number of turns j has to be increased. Conversely, to increase response to high frequencies, a low capacity coil (reduced number of turns) is required and sensor size should be small with respect to the wave length Moreover, to obtain satisfactory rejection of the electrical field, the coil is shielded which in turns limits frequency response via a stray capacity, The first type of sensor consists of a flat coil shielded with a split shield. Q =j!S EO Er. ds (2) where s is the surface area of the hemisphere. Taking into account the value of Er (l), the induced charge is : Q = ~01.f~Er ds = soi/3Ei ds cos 8 = 311r02EoEi (3) It can be observed that this charge is three times that of a disk with a radius ro. The voltage Ve at the input of the ampli-, fier depends on this charge, Q, as well as on the capacitance between the hemisphere and the sphere and the input capacitance : Ve = Q ChtCc: 31Ir20scEi -= Ch+Ce (4) Under these conditions, the effective height par hemisphere is : Ve heff l/2= Ei 3dso = Ch+Ce (5) ThistYpe of device can be obtimized to obtain satisfactory sensitivity to low frequencies (large diameter, large number of turns') or response to high frequencies (small diameter, small number of turns). The second type of sensor is a Moebiusloop formed by two half loops made of coaxial cable 4 A4 Since the coaxial structure is adapted at output, the limitation at high freauencies does not show up as long as the wavelength remains significantly longer than the diameter. Sensitivity to low frequencies corresponds to that of a frame of the same dimensions with two turns, therefore poor. The approach adopted consists in making the coil by using a coaxial line. The outer conductor (shielding) is discontinued at each turn to prevent formation of short circuit loops. The effect of the break in impedance due to the splits in the outer conductor is compensated for by a corresponding adaptation which is obtained by connecting a network, corresponding to the characteristic impedance of the line used for the coil. To increase this device's sensitivity, while maintaining compactness, a magnetic core is place inside the coil. 4. - PARTICULARITIES OF THE TRANSMISSION CHANNEL. 4.1 - CONNECTORS The required passband of 100 MHz for lengths up to 300 m lead to usinggrad indexe fibers (100/140 urn). Although actual technology enables manUfacturing connectors with low optical lOSS (1 to 1.5 dB) the use of optical connectors on the worksite requires particular care at each connection-disconnection operation to avoid optical attenuation with the resulting decrease in cfynamic range. To prevent this derating and obtain a constant transfer function, the optical elements and their associated electronic circuits are enclosed in the electrical connectors. All the connection-disconnection operations are thus performed electrically. This dual sub-channel device with electrical input-output comprises a bidirectional optical modem. 4.2 -TRANSMISSION This type of sensor provides a maximum sensitivity of + 1 v/m ( S/N >40 dB ) for a bandwidth of TkHz to 150 MHz. 3.- OPTICAL TRANSMISSION PRINCIPLE The transmission line provides two channels : -a telecommand channel routing the various commands to the sensors (switching on, range switching, calibration, etc..), -a signal channel transmitting information from the sensors as well as service signals (calibration, battery, condition, etc..). The telecommand channel has a restricted passband (50 kHz) and makes use of frequency coding of commands as well as analog storage of signals (memory capacity) in order to obtain maximum immunity (operation in ambient electric fields up to severa hundred kV/m). The signal channel should have : -a transfer function independent of temperature, ageing, condition of the connectors and the optical fiber, -a wide passband (100 Hz - 100 MHz), -satisfactory linearity (2%), -a wide dynamic range (50 dB). Electrical/optical conversion can be performed by a laser diode or a LED. The laser diode has a passband and a power input about ten times higher than that of the LED (500 MHz against 50 MHz and 2 mW against 200 uW) *However, it is not very stable in temperature and requires thermalservo-control of the transmitted power). The LED was finally selected since its association with a compensation circuit enables considerable extension of its passband. Frequency compensation is obtained by progressively increasing the gain at higher frequencies. However, this operation requires : -a sufficiently fast diode, -a slow decreasing slope, -a passband independent of the level and temperature up to at least 150 MHz. These conditions are fulfilled by a LED specially developed for this application 50 pm chip and micro-lens). Figure 7 below indicates the frequency response for this type of diode before and after frequency compensation. The passband at -3 dB elec. is increased from 50 MHz to 150 MHz. ? the LED used in the optical line -._-.__ before compensation _------ after compensation. - 22 Figure 8 - Shows'the compensation.circuit diagram.. Figure 8 - Principal of the LED frequency compensation circuit. 4.3 - RECEPTION The optical electrical conversion can be performed by an avalanche orPIN photodiode. The avalanche photodiode has a gain of 10 to 100 times that of the PIN photodiode. However, it is very sensitive to the temperature and requires a high supply voltage (100 to 300 V). The thermal instability of the avalanche photodiode can be corrected by a servocontrol device.Its high gain enables building sensitive receivers with a low input impedance However, the high dependence between the junction temperature of avalanche diodes and their gain results in a non-linear thermal capacity effect deforming the low frequency signals (100 kHz). This deformation shows as a non-linear differentiation effect (fig.9). Figure 9 - Deformation of an AF signal produced by the thermal effect of the junction of an avalanche diode For these reasons , preference was given to a PIN photodiode with a low junction capacitance associated with an impedance matching amplifier with a very high negative feedback resistor (20 kQ )(figure 10). - This device applies a reference voltage to the input of the optical line, detects the level received by the optical receiver and, after A/D conversion and linear/log transcoding, commands a set of attenuators. It has the advantage of an extended passband, which is independent of the attenuator position, and a total absence of distortion. 6.- DIGITIZING The digitizing system developed for acquisition of EMP data enables : -high-speed sampling (2 ns) -a large number of samples (5000) -8 bits precision. 6.1 - OPERATING PRINCIPLE Actual technology provides no simple means of sampling followed by digitizing at a speed of 2 ns. This digitizing is obtained by using 50 samplinq channels which are scanned sequen- Figure 111- Block diagram of the high-speed digitizer Each sampling head thus works at a lower speed of 50 x 2 = 100 ns (figure 12). loons 100 s Figure 10 - Prinzple of the optical receiver This very high negative-feedback resistor, after solving certain technical difficulties (necessity of an unwanted capacitance 0.50 pF across the resistor) enables obtaining a dynamic range of 50 dB (optical power 30 VW) while preserving a passband of 150 MHz. The transmitter-receiver assembly associated with a grad indexe fiber of 100/140 urnenables building a 100 Hz - 100 MHz/50 dB optical line over 300 m. 5. - OPTICAL LINK PROCESSOR The optical link processor enables telecommand of the sensors via the optical link and pre-processing of the signal before digitizing. This pre-processing consists of : -calibration of the optical link with automatic adjustment of the transfer function, -filtering of the signal as a function of the required passband and the sampling rate speed, -matching the output level to the digitize input level in order to optimize the dynamic range. The constant transfer function is obtained, by an AGC device using a calibration signal. Figure 12 - Distribution in time of the instants at which the various channels are w. After sampling, the signal is stored in analog memory (100 memory elements per channel) The process is continuous and after 50 x 100 samples (time = 100 ns x 10 = 10~s ) a part of the old information is lost and is replaced by new information (drum memory) (figure 13). When the equipment is triggered, the sampling procedure is stopped, the 50 analog memories are read and the A/D conversion is performed at a slow rate. The continuous sampling before triggering enables providing information of events before as well as after the instant of triggering. The digitizer is controlled by a microprocessor which, in particular, performs the corrections required by the inaccuracy of the sampling heads and the analog memories. - 23 6.2 - PARTICULARITIES OF THE DIGITIZER The diqitizer has three main properties adapted to EMP meaSUrement : -drum memory associated with a System of Pre and post triggering, -very high digitizing rate (2 ns) I -large number of points (5000). The drum memory and the pre and post triggering system enable digitizing signals whose exact shape and polarity are unknown.Figuras 13 & 14 givean example of a signal with a positive and then a negative half-cycle, produced by equipment adjusted for negative synchronization but with pre-triggering. The high sampling rate associated with a large number of points enables digitizing signals which simultaneously have high speed and long duration (signals occuring on long transmission lines with protecting elements limiting the signal) (figure 15 and 16). For this type of signal, the digitizer provides a 150 MHz analysis passband for 5000 x 2 ns = 10 us. FinalLy, the large analysis window enables acceptence of signals occurring at different moments without using delay lines. 7. - DATA PROCESSING SYSTEM The coreof the data processing system is the IBM-PC, version AT, with disk drive (20 Mbytes) and floppy disk drive (1.2 Mbytes), color monitor, graphic printer and color x, y plotter. The software consists of 6 modules : -device management -results filing (raw or after pre-processing) -pre-processing of results -results display (raw or pre-processed) -firing aid -mathematical processing. 7.1 - Device Management This module ensures remote control of the digitizers and sensors and setting of the optical processors connected to each optical line. It performs all the channel calibration operations and monitors system operation (battery status, internal tests, etc). 7.2 - Results Filing This module enables two results filing modes : on disk (temporary) and on Flo.ppy disk (permanent filing). It also enables accurate description of the test conditions : date, time, composition of each system, position and direction of each sensor. 7.3 - Results Pre-processing Pre-processing of results enables shifting the signals to a common time origin and reduction, with operator consent, of the number of points for each measurement, if the signal width does notwarrant filing of all the points. 7.4 - Results display This module enables display of one or several signals ; this display facilitates correlations. In particular, simultaneous display of the incident fields and inducted currents and voltages enablesa betterundertanding of the induction processes. - 4A4 In the same vein, simultaneous display from a given of the fields obtainedviewed point and with incident fields of different values enables clear identification of the system's nonlinearities. Finally, comparison of the fields obtained from neighboring sensors enables detection of operating incidents. 1.5 - Firing Aid This module assists the operator in setting the various parameters in the acquisition systems : -sensitivity (sensor) -filtering frequency (optical processor) -sampling rate ) ) digitizer -off-set -pre or post triggering ) -etc. One to three fixings are necessary to reach an Optimal setting. This relatively rapid setting procedure is possible due to the sensor's high in&antaneous dynamic range 1>50 dB). The dynamic range, along with the switched dynamic range (70 dB) allows measurement Over a dynamic range of 120 dB. Example : Electrical field sensor (ranges : + 3kV/m, + lkV/m, 5 38@J/m, + lOOV/m, t 3OV/m, T lOV/m, 5 3V/m, $- 1 V/m). For the first firing, the SenSOl iS Set On a lesssensitive range (3 kV/m). If after digitizing, signal amplitude is greater than 3kV/m the program provides the option to the operator to change sensors or to decrease the incident field. If the field is included between 3kV/m and 30 V/m, the program suggests setting the sensor to the nearest range. If the field is less than 30 V/m, the sensor is set at this level of sensitivity and another firing is started. The new value obtained is used for resetting the sensor. If the range selected is the most sensitive and the signal amplitude is still not adequate, the program suggests either to increase the incident field (if the other sensors or installation allows it)or to improve the dynamic range by using an 80, 40, 20 OX 10 MHz filter (gain of 3 to 12 dB). Hence, with a 20 MHz filter is it possible to detect signals starting at 5 mV/m. If the computer controlling data acquisition is linked to the pulse generator delivering the signal, setting the range in function of generator voltage is easier. In the same manner the operator can set the limits for the ranges by restricting the measuring dynamic range. A second important setting is selection of the filtering frequency and sampling rate best suited to the measurement. These selections are made after determining, fwom the fastest transition, the passband of the signal under analysis. Based on this value, the filter which is best adapted to the measurement (maximum of dynamic range) and corresponding sampling rate is used. This selection is particularly of interest when examination of long events (low frequency resonance or computer sequences) require a large analysis window and slow sampling rates, - 24 - hence, a risk of undersampling in the absence of an adequate filter. 7.6 - Mathematical Processing This model enables calculation of a certain number of mathematical functions (FFT, correlation, powers, etc.). When voltage sensors are used and if circuit impedance is knowen,the module allows going from voltage to current via a Laplace transformation. 8. - CONCLUSIONS The simultaneous utilization of active sensors with high sensitivity and a large dynamic range, an optical transmission line with a wide band and stable transfer function, and solid state high-speed digitizers with a large measuring time range is perfectly suited to the measurements required by the EMP simulator. The associated date processing system also enables maximum automatizing of the fire control system while leaving the operator the final decision, but after having provided him with the elements for that decision. 1~ 5!LA I Figure 16. Digitizing in the presence of fast signals with a long the duration (2) Figure 13 -Figure 17, EMP data acquisition system block diagram. Figure 14.Principle of the drum memory. (2) Figure 18. Main Sensors - - BETWEEN 25 5 - THE INTERFACE ESD TESTING: SIMULATOR AND EQUIPMENT UNDER P. Richman KeyTek Corporation Massachusetts, Crucial issues regarding the interface between the ESD simulator and the Equipment Under Test (EUT) include discharge current peaks that are vastly different from simply-calculated Values, and failures of the EUT at both low and high, but not intermediate voltage These phenomena can be exlevels. plained and mathematically modeled in terms of circuit inductance and freeThe more inclusive space capacitance. circuit model that results, gives significantly improved agreement between calculated and experimental electrostatic-discharge current waves. TEST and A. Tasker Instrument Burlington, Bd U.S.A. a hand-held metal object key, bracelet or ring. like a tool, Values called for by various Standards and used by individual organizations range from 60 to 300 pfd for C, and from 10 to 10,000 ohms for R (l-5). ESD ,- EUT Introduction The peak current that flows during an electrostatic discharge (ESD) from an ESD simulator can be vastly different from the value intuition might lead one to expect. It can be at least as low as one-tenth, or at least as high as ten times, the value computed by dividing stored -- or test -- voltage by the simulator's nominal internal resistance In addition, the discharge current waveform in both simulator and actual human-body discharges often bears little relation to the simple, single R-C equivalent circuit in widespread use (l-5). GROUND Fig. 1: Conventional, for Personnel Discharge PLANE Single R-C Model Electrostatic Fig. 2 shows discharge current due to a typical human-body discharge from a hand-held metal object. Instrumentation for converting the discharge current into a voltage suitable for oscilloscope monitoring was built as per reference (1); oscilloscope bandwidth was 400MHz. Even though the circuit The two factors most responsible for these often huge discrepancies are circuit inductance and capacitance to free space. The conventional model for personnel electrostatic discharge is given in Fig. 1. It consists of a simple capacitor C, charged to voltage V, and discharging into the victim equipment -the EUT or Equipment Under Test -through resistor R. The "low" end of C is most often connected to a ground plane or to a point on the EUT, or both. A discharge tip, connected to the resistor R, is advanced toward the EUT until an arc occurs, simulating the spark that leaps from a finger or from Fig. 2: Typical @SD Current Discharge Wave from a Hand-Held Metal Object. (Steep-rise edges retouched for readability) 5kV Initial Charge Level 2.5A/half cm, 2ns/half cm - model of Fig. 1 is in common use, there is simply no way in which it can begin to account for the real-world Fig. 2 waveform, specifically for the sharp, high-amplitude initial spike. (Others have also reported initial spikes (6).) The single R-C model of Fig. 1 is inadequate in that it ignores: 1. The human body and/or ESD simulator circuit inductance, which ranges from 0.5 to 2 uH. 2. The 3 to 10 pfd, almost inductancefree capacitance to free space of the human hand. 3. The typically 5 to 20 pfd, almost inductance-free capacitance to free space of the victim EUT itself. Circuit Inductance Reference (l), an IEC draft ESD standard for Process Control, specifies a one-meter long ground return of 20 mm width. However for calibration purposes, the same draft standard calls for a discharge circuit, including the ground connection, that is "as short as possible". Calculations, confirmed by tests, give total circuit inductance including internal simulator circuitry as well as the ground return itself, of about 1.7 yH for the one-meter ground return. Similarly, a figure of 0.7 uH results for a typical R-C network with a calibration-length ground, with a length on the order of 30 to 40 cm. 26 - Fig. 3 shows the addition of total circuit inductance L to the simpler circuit of Fig. 1. Table 1 shows the large effect that L can have on network "efficiency" v\ , defined as the ratio of peak current Ip during discharge, to the "intuitive" peak of V/R; multiplied by 100, to obtain per cent. (Efficiency without L must be lOO%.) Calculated values of I were computer-derived from appropria! e solutions to the series R-L-C circuit of Fig. 3, and were spot-checked via experiment. I was calculated for a stored voltage o!? 5kV, but for different voltages the values of Ip can be scaled proportionately; ignoring preionization and other effects. R and C values come from representative ESD test Standards, as listed in Table .te;fy v - TIP 1 RETURN, 1 GROUND PLANE Fig. 3: ESD Model Modified to Include Total Circuit Inductance, L Table 1 R and C Values for the R-L-C Equivalent Circuit of Fig. 3, with Efficiency hgiven for Realistic Ground Return Inductance, 1.7 uH Ip is peak current for a stored voltage of 5kV tp is time of occurrence of Ip Standard Organiza- or Draft tion (l-5) Standard - LP Q. =lOO,lR (p:d) (ohms) --- (%) 150 64 Ip for 5kV (A) tP ns 21 18 1. IEC(1) 65 (Seer) 80 (Draft) 150 2. MIL(2) 883 B 100 1,500 97 3.2 5.6 3. NEMA(3) Part DC33 (Draft) 100 1,500 97 3.2 5.6 4A. EIA(4) PN-1361 (Draft) 100 500 87 9 4B. EIA(4) 1, 60 10,000 100 .5 1.4 5. SAE(5) 51211 300 5,000 100 1.0 2.9 60 10 6 6. Cart simulation 28 10 1.6 561 27 - Inductance L was taken as 1.7 uH, 1. representative of a typical simulator including a 20 mm wide ground return of about one meter, the length recommended by the IEC draft standard. Note the vast differences, particularly for the IEC 150pf/150fi network, between V/R (33A for 5kV) and the Calculated value, 21A, for peak current Ip for 1.7 PH circuit inductance. For the "calibration" circuit inductance of 0.7 uH, the same 150pf/150n IEC network gives a calculated peak current of 25A. Typically, arc and corona effects reduce this still further, by as much as 20 to 30%. Thus a 5kV stored voltage with a 150 ohm resistor will typically result in a peak current of only 16 to 18A. "Calibration" in test laboratories may report defective simulators, with only one-half required output! (The IEC specifies a peak current of 50% to 90% of stored voltage divided by resistance, thereby covering the situation quite completely. Unfortunately many calibration laboratories simply calculate V/R, and either ignore the IEC specification or neglect to calculate the effects of even the 0.7 uH "calibration" inductance.) Neither peak current Ip nor peak time tP respond to network differences in a simply proportional way. The effect of changing from 150 pfd/l50 ohms to 60 pfd/lO ohms, for example, is rather small; current peak increases from 21 to 28~, time to peak decreases from 1B to 16 ns. Yet the nominal "efficiencies" of the two networks differ by over an order of magnitude. The explanation is that inductance is the controlling factor. Until the simulation circuit resistance gets large -- 500 to 1500 ohms -- or more accurately until network efficiency exceeds go or g5%, circuit inductance dominates performance. Capacitance to Free Space; Interaction with Inductance Every object has capacitance to free space -- or to the walls, floor and ceiling of the room in which it is located. For a spherical object of diameter dl and a room (also taken as spherical, for simplicity) of diameter d29 the capacitance is given by reference (7) as: C=O.556 x Kdld2/(d2 - dl) in which dland d2 are in cm, and Km for air. (1) 1 For a human hand or arm in a room of typical dimensions, the term d2/(d2-dl) approaches unity, so that c- 0.556 dl (2) For a hand of approximately 9 cm "diameter", capacitance is thus on the order of 5 pfd. Note that this capacitance is almost inductance-free. The inductance of a finger, hand and/or forearm may be calculated from reference (7) as: L=O.O021 [2.303 loglO(4j/d-11 uH (3) in whichQ and d are length and diameter, respectively, of the finger, hand or forearm; again in cm. Table 2 gives results, along with approximate values of capacitance to free space, for all body segments involved. Use of a hand with key has been assumed, as this is rapidly becoming a de-facto standard for worst-case ESD simulation. It represents an ESD event involving a handheld metal object such as a tool, ring, bracelet, or indeed an actual key. Table 2 Approximate Dimensions, Estimated Capacitance and Estimated Inductance for Various Sections of the Human Body (d = diameter,&= length) d cm -- Q.c L cm pf' --- PH Fingers holding key 2 6 2 .02 Entire hand holding key (to wrist) 7.5 12.5 5 .02 Forearm (wrist to elbow) 9 30 10 .l Full arm (wrist to shoulder) 9 60 20 .27 Torso (shoulder to waist) 30 60 20 .13 Whole body (torso plus lower body) 30 120 40 .43 Computer solutions are given in Table 3 for peak current I and peak time tp, from the differentia P equations that describe performance of the circuit of Fig. 3. Solutions are given for values representative of appropriate combinations of the hand and arm from Table 2, using a compromise inductance value of 0.1 PH. Solutions are also given in Table 3 for the R-C values specified in various standards as set forth in Table 1, for both "calibration" (0.7 uH) and normal l-meter (1.7 uH) inductances. A resistance of 200 ohms is used for the small capacitance val- - 28 - Table 3 Computed Values of Peak Current Ip and Peak Time tp Ip computed for 5kV; simply scale for other voltages (For virtually all parameter combinations except R=lOK, risetime,T, lies between 25% and 65% of peak time tp.) C (Pfri) R (ohms) c& I, (amperes) for bandwidth= Infi400 60 100 MHz nite MHz MHz 50 50 200 :1 .1 10 19 16 6 15 2.4 14 ;:", 10 200 200 1 :1 18 18 16 17 60 10,000 .5 ; 7.5 100 100 150 300 500 1,500 150 5,000 1 .7 1.7 .5 .5 ; .7 1.7 ;:: .7 1.7 .7 1.7 25 21 1.0 1.0 1.0 1.0 ues that simulate the hand and arm, but is not a major determinant in Ip or tp, over a wide range of resistance values. It is assumed that the victim EUT has significant capacitance to free space in the surface area immediately adjacent to the point of ESD application; i.e., it is another, larger, and also virtually inductance-free capacitance. From reference (7), for example, the capacitance of a 30 to 40 cm diameter disc to free space can be calculated as 12 to 15 pfd. This might represent that portion of a victim EUT panel or keyboard at whose center the ESD was applied. Without such a "ground plane", free-space capacitance effects due to finger, hand and arm will be very much lower, due to the lower total circuit capacitance that will result. (EUT "ground plane" capacitance is effectively in series with hand capacitance.) In this connection it is worth noting that the IEC-designed coaxial "target" (1) performs far better in making current-spike measurements when it is mounted to a ground plane on the order of 40 x 40 cm. In add tion to the "infinite bandwidth" theoretical values given in 1: ;:: 25 24 21 21 1.0 1.0 1.0 1.0 for bandwidth= 400 loo 60 MHz _- MHz - MHz - .4 .6 .7 .7 :; 1.0 1.2 1.2 1.8 1.3 2.0 1.0 1.1 1.4 1.5 2.1 2.4 2.4 1.4 ;:: z-2 . :: 5.2 10.1 5.7 10.6 ::: 2:: 5:; . 10 11 .7 1.7 tn_(ns) Infinite .6 10 18 1.3 2.9 10 18 3.5 4.0 8 12 7.9 9.1 2.8 10 14 11 12 12 20 13 21 11 17 11 17 Table 3 for Ip and tp, computer solutions are also included in the table for the same waveforms viewed with oscilloscopes of finite bandwidths: specifically 400 MHz, 100 MHz and 60 MHz. Data in Table 3 go a long way towards explaining differences between measurements made by different investigators. Simulations with 60 pfd and lOK, for example, will be vastly different depending on the simulation capacitor's physical size, and on whether the simulation resistor is 12 cm long -hence not simulating a finger/hand combination -- or short, and contained within a metal enclosure to which it might have, for example, .5 ofd stray capacitance. For 6Opfd/lOK, Ip is .5A at 5kV. But if stray capacitance -- or capacitance of the simulating 60 pfd to free space -- is considered and a 400 MHz scope used, then from Table 3, I will be 6A for an arc resistance of !ZO n; and 10A with infinite oscilloscope bandwidth. Yet the value shown with 60 to 100 MHz instrumentation will range from only 1.5 to 2.4A. And after all, the 0.5 pfd stray is only 10% of the 5 pfd representative of the human hand -- which at 400 MHz gives 16A for 2OOn, as shown in the table. - Thus all of the simulation circuits in references (1) through (5) miss the the hand/forearm combination point: has a 5 to I.5 pfd capacitance to free space, and it is coupled to the discharge arc with only 0.05 to 0.1 uH. The result is a super-fast edged, short-duration (1 to 4 ns) spike of 15 to 30A, for a stored voltage of Only 5kV. Experience shows this spike can be crucial in causing EUT malfunction, but it is neglected by all existing standards. And if it is accidentally viewed on an oscilloscope, its amplitude is typically underestimated by a factor between 2 and 4 by the 60 to 100 MHz instrumentation in common use. Any saving grace that a simple R-C ESD simulator may have is that the simulation capacitor can itself have a capacitance to free space! This accounts for the sharp wave-start so frequently seen in simulator current waves. Fig. 4 shows a typical case, for the EIA values of 100 pfd and 500 ohms (4). But this is a far cry from the sharp, 1 to 4 ns spike of 15 to 30A peak (for a stored voltage of 5kV) that is generated by a hand-held key, as shown in Fig. 2. ESD simulator's discharge current output, the amplitude of the sharp initial edge generated by the simulator capacitor's own capacitance to free space, is usually less than peak current due to the simulator's basic R-C. For this reason it has been seen as merely an unpleasant anomaly in the wave, due to "parasitics". In point of fact, it provides whatever inadequate sharp-risetime "punch" such simulator waves do have. 29 - 5Bl Corona effects at higher voltages -above 3 to 6 kV, depending on discharge tip geometry -- reduce the sharpness of the initial spike or step. This effect most likely accounts for the fact that many equipments that can pass ESD tests at 10 kV, say, at which level corona has seriously reduced risetime, will fail at only 5kV, due to the steep initial rise of the spike or step. At voltages of 15 to 20 kV, failures may start again, as the sheer magnitude of the di/dt, even with heavy corona, once again becomes high enough to cause equipment malfunctions. Fig. 6 shows computer-generated current discharge waves for the Dual R-L-C circuit of Fig. 5, both without (6a) and with (6b) simulated arc oscillations. Hand-simulation values are 7.5 pf, 200 n and 0.1 yH. Body-simulation values are 100 pfd, 5OOfi and 0.7 PH. Fig. 6b corresponds well with the human-discharge current of Fig. 2; i.e., the Dual RLC model works. In the typical The New, Fig. 4: Typical ESD Current Wave from a Single R-C ESD Simulator (100 pfd, 50051) (4). Sharp WaveStart is due to the Simulator Capacitor's own Capacitance to Free Space 5kV Initial Charge Voltage 2.5A/half cm; 2ns/half cm Dual RLC Circuit Model Fig. 5 shows the new, Dual RLC Circuit Model that seems to best replicate the effects of circuit inductance, plus hand capacitance, in an electrostatic discharge. RB 150-1500 LB .5-z DISCHARGE Two parallel R-L-C paths are provided; one for the body (CB, RB, LB), the other for the hand (CR, RR, LR). Since the impedance from Discharge Tip to ground during the discharge is generally low -- due to the victim EUT's own capacitance to free space -- the two R-L-C paths function as almost independent current sources. Thus the waves they generate are superimposed. CR/RR/ LH generates either the steep-rise initial spike, or for higher voltages at which corona effects cause pre-ionization, a large initial step. CB/RB/LB then generates the longer wave that follows, carrying the often less-damaging energy stored on whole-body capacitance. Fig. -- 5: The Dual RLC Circuit Model for ESD; Incorporating Separate, Parallel Paths for Body and Hand Discharge _ 30 - Fig. 7 shows a typical current-discharge wave from a practical ESD simu. later that was designed to reproduce the Dual R-L-C model of Fig. 5. It agrees well with both Figs. 2 and 6b. 0 Fig. 6: 4 8 1211s 0 4 8 12 ns Computer Solution for Discharge Current from the Dual R-L-C Circuit of Fig. 5; (a) Without and (b) With Superimposed, Simulated Arc Oscillations cH=7.5 pfd RH=2OOn LH=O.l uH CB=lOO pfd RB=5OGn LB'1.7 I-tH 5kV Initial Charge Level 5A/division, 4ns/division probably still causes a large proportion of ESD-simulation failures. 4. A parallel RLC/RLC circuit model (the "Dual RLC" model) gives excellent general agreement with initial edge and initial spike experimental data. It is quite practical to simulate 5. the Dual RLC model with physical components, while nevertheless retaining the convenient one-meter ground return. Data from such simulators agree well with both calculations and data from actual personnel electrostatic discharge. The Dual RLC model represents the situation well: CH discharges through a low inductance to give the initial spike simulating the human hand; CB discharges through a higher inductance, to simulate the longer wave that conveys energy stored on the entire body. 6. Simulators not incorporating the CHRH-LH hand-spike simulation path may well not be able to induce failures at 3 to 6 kV in the same way that actual personnel discharges can do. Thus ESDtesting with such instrumentation may not represent the reality the equipment under test will face when placed in service. References Cl1 International Electrotechnical Commission IEC 65(Secr)80 Draft Standard: Electrostatic Discharge (for Industrial Process Control). i-21 MIL STD 883B, Test Methods and Proceedures for Micro Electronics. Fig. 7: Actual Discharge Current from a Practical ESD Simulator Embodying the Dual R-L-C Circuit of (Steep-rise edges reFig. 5. touched for readability) 5kV Initial Charge Level 2.5A/half cm, 2ns/half cm c31 NEMA, Residential Controls, Envi- ronmental Testing for Electronic Controls, Part DC33, Proposed June 24-25, 1982. c41 EIA (Electronic Industries Associ- ation) Standards Project ~~-1361, Environmental and Safety Considerations for Voice Telephone Terminals, Draft 5, Nov. 24, 1981. Conclusions 1. Any ESD circuit model that doesn't include inductance can't simulate reality well enough for test purposes. c51 2. Capacitance of the hand to free space causes a spike at voltages to 5kV, and an initial fast edge at higher voltages. Both are grossly under-estimated by 60 to 100 MHz instrumentation, while measured adequately with instrumentation of 400 MHz and above. C61 King, W.M. and Reynolds, D., Per- The inevitable 0.25 to 0.5 pfd ca3. pacitance to free space of the simulator's internal capacitor is the only tie to fast-edge reality that many simulators have; and it is too small by a factor of at least ten. Nevertheless it SAE Standard Recommended Practice Information Report J-1211, June 1978, P 20.99. sonnel Electrostatic Discharge: Impulse Waveforms Resulting from ESD of Humans Directly and Through Small Hand-Held Metallic Objects Intervening in the Discharge Path, Proc. IEEE Int'l Symposium on EMC, Aug. 18-20, 1981, pp. 577-590. II71 Terman, F.E., Radio Engineers' Handbook, McGraw-Hill, 1943, pp 48, 113. 6 - 31 - RECENT OF COUPLING DEVELOPMENTS IN THE PATHS OF ESD THROUGH Michel UNDERSTANDING A METALLIC CABINET Mardiguian and Donald R.J. Don White Consultants, Inc. Gainesville, Virginia, USA Abstract White This also serves as a reference to further shielding effectiveness assessment. One must remember that the ESD, although quoted Based on recent studies made on ESD event statistics and ESD modelization for furniture and human discharge onto a computer frame, in-depth analysis is made of ESD current routes on a metallic frame and mechanisms of re-radiation inside equipment. Although a metal box should behave as an efficient shield, it k shown that this does not happen because high-frequency spectrum (up to the Gigahertz region) of ESD excites all existing seams and slot leakages. Measured values of E and H fields (inside the cabinet) near the discharge area and some peculiar aspects, like the effect of a discharge on a screw protruding significantly inside, arc discussed. The Electric and Magnetic field amplitudes lead to some discussion on the wave impedance of the ESD reradiation inside the box and its near field/far field transition. This, in turn, allows a better prediction of the noise voltages induced in nearby PCB traces or flat cables. Grounded Metal Plate Electric 01 Magnelic Field Probe - Ground Plane E \ \ Spectrum Background _____- Figure In the past few years, significant progress has been accomplished in understanding the electrostatic build-up and discharge mechanisms, their simulation and the ESD hardening of integrated circuits. Several computer manufacturers and independent experts have disclosed the results of their studies. However, in contrast with the large amount of data on human body and furniture voltages, capacitances and resistances, discharge rise times and waveforms, etc., relatively few quantitative studies have been done on field amplitudes around an ESD discharge [ 1,2,3]. This paper is a follow-up of a study of Ref. 4 which examined sequentially the mechanisms of the ESD coupling via equipment cabinets and external cables up to the distributed victim circuit. Here now is an attempt to quantitatively evaluate the Electric and Magnetic fields near an ESD discharge path, and the factor influencing field re-radiation inside a typical electronic enclosure. In these tests, the electrostatic discharge was simulated using a Schaffner NSG 430 (150 Q, 150 pf Network). The E and H fields were measured by miniature monopoles, short balanced dipole and magnetic loop (electrically shielded). The probes were connected to an Electra-Metrics ESA 1000 Spectrum Analyzer. A slow scan speed and sufficient RF attenuation were chosen to avoid Spectrum Analyzer error due to the broadband nature of the measurement. E and H Fields Values and Polarization, Discharge to a Vertical Structure, B2 from Over a Conductive an ESD Ground First, the test set-up of Fig. 1 has been arranged to measure the field amplitude facing to an ESD “Zap,” in the absence of any protective shield. This would be the case of a discharge to a metal object near an equipment having only a plastic enclosure. l-Experimental Analyzer set-up for ESD field measurement “static,” is certainly anything but a static phenomena: Within few nanoseconds, a localized electric field of several kilovolts per cm (corresponding to several hundred kilovolts/meter) collapses to zero while in the same time, a localized magnetic induction raises up to several Gauss! With a scaling factor (a few amperes discharge instead of ten kilo amperes, and a spectrum of few hundred MHz instead of few hundred kilohertz) the ESD is in fact a miniature version of a lightning stroke. Few documents have stressed this fact [ 1 & 41 and others have reported field strength values. These values were generally measured at one meter. Although the non-uniformity of the field makes closer measurements less accurate, during our study E and H field magnitudes have been measured at 10 cm, 30 cm and 1 meter. The vertical structure was a 50 cm by 6 cm aluminum plate, firmly bonded to the copper ground plane. The ESD gun was set to 10 kV and an arc discharge, with a slow repetition rate was made on the upper tip of the plate. The ground return for the ESD gun was a flat strap about 30 cm long to avoid the possible influence of both inductance and location of the return conductor. For the same repeatability reason, the orientation of this strap was kept always in the vertical plane formed by the gun and the structure which was discharged upon. Figs. 2a, 2b, and 2c show the results of electric and magnetic fields, after bandwidth and antenna factor correction. A few remarks are in order: a) The 1 meter results correlate within few other reported measurements, discharge simulators about done + 15dB with with similar b) Compared to the I meter results, the 30cm and 1Ocm results seem to show a (distance)W”2 dependancy instead - of a (distance)-2 or (distance)’ as one would least in the induction (near-field) region. expect, 32 - at structure, the ESD generates a predominately magnetic (low impedance) field in the induction region, tending to a 120 x ohms wave impedance in the far field zone. Since the change over of near to far field is wavelength dependant, the transition occurs at different frequencies for the various distances of the experiment; the change is very pronounced for the IOcm case. In this experience, it must be reminded that the radiator is the whole circuit formed by the simulator and Frequency 1 3 10 I” MHz 30 100 300 3MH7 1 30MH/ 1 IOOMHI 1 300MH/ 1 5OOMHr ~~ m men a ion u Table l-Average .m 1 3 10 30 Ftequency Figure 2a-ESD Field at 1 meter 10 300 Free Field Radiation Frequency 3 100 in MHz 30 in MHz 100 300 120 120 sz 100 100 z. 9 % 80 5 80 g D s N : 60 60 g 40 40 g d 20 20 m’ Q N N 3 10 30 Frequency Figure 2b-ESD 100 P m I” Wave Impedances of ESD Radiated Fleids The statement that the field is predominately magnetic near the discharge path may be surprising. There is a common belief that ESD, “being electrostatic, has to be an electric field.” A close look at the discharge network can clarify this: Simple Field theory says that in near field region, low impedance (<337B) sources will radiate predominately magnetic fields, while high impedance (>377Q) sources radiate predominately electric fields. The ESD simulator used follows the IEC-65 recommendation, with an internal resistance of 15OQ. Therefore,it behaves more as a magnetic source in near field. Will “real-life” electro-static discharges really appear like this? In the opinion of the author, the 15OQ value is a fair compromise, but it has the drawbacks of every compromise. Actual furniture-type discharges 161 from large metal objects, carts, chairs, etc. may exhibit dynamic impedances IO times smaller or even less, creating more magnetic field in the near region. While human body resistance, being at least 10 times higher, will create less magnetic field. The IEC-65 somewhat makes up for this discrepancy by recommending such voltages (8Kv and 15Kv) that they force a current about similar to a furniture discharge current. The possible effect of wave impedance on victim circuits exposed to the ESD radiated field will be discussed in the next section. 300 in MHz Fields at 30 cm Frequency 10 30 I” MHz 100 d) A rough integration of the electric field spectrum over the frequency domain gives the following approximation for its time-domain peak value: at 1 meter = 70 volts/meter at 30 cm = 120 volts/meter at 10 cm = 220 volts/meter 300 100 N : 80 3 Voltages ._____~ Induced in Nearby Printed Circuit -___ ___-. and Other Small Circuits Boards -.__ .._ m” 60 D 10 Figure 30 100 Frequency ,n MHz In the second part of the experiment, the antennas were replaced by a PCB having a unique trace representing a loop of 1Ocm x 10cm. This trace was alternatively terminated into 1 kilohm, open-ended, then terminated into a short. The voltage picked-up was read on the spectrum analyzer, with all precautions to prevent possible pick-up by the coaxial cable. 300 2c-ESD Field at IO cm the vertical discharging structure. Seen from an antenna located at 3OOB above 300MHz --@ 1Ocm from about 1OD around 1OMHz to 500 above 300MHz. Fig. 3 shows the induced voltages being oriented tangent to wave front. that for actual arcing on a metallic the PCB The effect of varying the far end terminating resistances is interesting in the prospective of understanding which of H field or E field coupling predominates. Fig. 4 shows the traditional model for a small rectangular circuit illuminated by an electromagnetic field. The E field creates a transverse voltage V2, which appears as a higb impedance source (current source) with an open circuit voltage: V2 = E x 2L x h x This indicates in dBpV/MHz, T cos 0 cos o! x (1) - 110 1 I Frequency in MIir 10 30 100 3 33 300 90 80 80 70 70 60 60 like the coupling l/F as evident coefficient on Fig. 2, while 10 MHz, in Eq. (3) starts to create series insertion 150-200 MHz, available ing the available voltage By comparison, the wir- loss, caus- at the 5OQ end. Finally, the H field spectrum above itself decreases like I/F’, to collapse B2 at the same time increases like F. Above ing less and less voltage 100 90 decreases ing impedance 110 100 6 - even more caus- rapidly. the values for magnetically induced voltage using Eq. (2) and (3), or the graphical method of Ref. (5) are shown also on Fig. 3. They are in fair agreement with the measured data. Curves Frequency I” MHz ! 1 3 10 30 100 I 300 A and B of Fig. 3 correspond on far end, which minimize left is the electrical contribution. is one order thermore C, Calculaird 0 Calciilated voltage based on ti Field coupling voltage based on E-Field couplmg the only only of magnitude supports “standard” Figure 3-Broadband voltage induced rn a 100 cm:‘PCB run located 10 cm from the ESD path, parallel to wave front (PCB not oriented for maxlmum H-Field interception). less then the magnetic is, Then, the PCB was rotated wave front, such as to intercept magnetic contribution termination contribution. 0 = angle between the E field plane of the loop and the direction trum the SOQ impedance analyzer) a high impedance will double the available will it. nullify On the other VI appearing and the direction of the “victim” receptor end the H field as a low impedance voltage, while 1kQ a shorted creates a longitudinal source of of that in field, near one. This fur- the radiation of predominantly 90” to be perpendicular the maximum magnetic case. In this set-up is so pronounced IOOOQ (curve A) to the flux. The in exactly the however, the that even with a far end it overrides the electric (voltage 60 of Frequency A B C end (spec- on the far end, like transverse is in a small circuit u = angle between propagation. tiiven statement on Fig. 5 and can be interpreted same way as for the previous Where What It is clear that this contribution the previous discharge results are shown induced contribution. magnetic. c Figure 4-TradItional model for voltages illuminated by an EM field. to a high impedance the magnetic or in all cases, wlh Ziar end = m (wen) with Zlai end”0 (short) z,,,, end (Receptor)=50 I1 I m end in MHz with Zlar end = 1 kll Figure B-Broadband voltage Induced In a 100 cm;‘PCB run located 10 cm from the ESD path perpindicular to wave front (PCB Intercepting maximum magnetic field). voltage source) with a value: In fact the difference MHz correspond between approximately curves (A) and (C) below to the ratio of IO lOOOn to 5OQ termination. Interestingly there the influence voltage vx = of varying impedances Vx across the receptor VI,,, is totally different. The end is: corresponds Effect Z, ,,,,(,,I,,, il) -~ ~--.--------~----1, t & + z,,..,,,,, (3) -= 5On Z ,,,,,,,,.=O.l A high magnetically impedance induced Z, voltage. n + .iw x 0.4~H on the A shor/ far end on /he fur nullify the Metallic by the normal Cabinet shielding on ESD Radiation ___ ____~_ the cabinet metallized) of curve (C) 15 volts. housing the ESD field the electronic should effect of the material. for the external has been discussed in a former pick-up paper by I/O well protected. real life enclosures are full of slots, scams, apertures, disrupt integrity. the shield cables, which [4], it seems that boards and wiring should be fairly be at- Therefore, internal However, etc., which end will max. iUli,lcJ il. Every shield discontinuity across the ESD current path will “shine” inside, with an efficiency proportional to its length com- On Fig. shorted. (or the spectrum of about and not accounting circuit will of a Typical is metallic tenuated Z, integrating If, instead of being plastic, circuit in our experiment, enough, to a peak voltage 3 the curve Therefore, (C), corresponds the electrical contribution to the far end being is minimum while the magnetic contribution is enhanced. The flat portion of the voltage spectrum corresponds to the domain where the H field pared to the half to and above wave length. 500 MHz, hibit significant leakage. mechanism, the “witness” The ESD spectrum any slot longer than extending up a few cm wilt ex- To show this effect on the ESD PCB was placed inside in lmm thick - aluminum rack; to calibrate the experiment, all mating surfaces have been thoroughly brushed and screwed and the ESD gun, set to IOkVolts was discharged on all sides and especially in the seam areas. No value exceeded the sensitivity level of the test set up. Then, several typical shield imperfections have been introduced by removing some of the top cover screws and inserting Imm cardboard liners under the seams to simulate an ungasketed cover with ordinary manufacturing tolerances. Fig. 6 shows the results. The induced voltage could be read up to 100 MHz, for the enhanced magnetic coupling (far end termination shorted) which demonstrates two points: a) the thin seam unequivocally spoils the protection by the box to ESD coupling b) the re-radiated magnetic. field inside is, once again, offered predominantly An interesting effect was also simulated: One of the threaded holes used to attach the top cover has been painted and the cover was mounted using a long screw, protuding about 2.5cm (I”) inside the box. The results are also shown on Fig. 6. It seems that the screw generates a secondary arc inside, between the fillets and the inner box surface. Frequency I” MHz 100 90 : E 90 N 80 : 80 s 70 7ov 60 60 m" u - Discussion -_____ m" It has been shown that a circuit illuminated by a typical IOkVolts ESD discharge is exposed to field values in excess of 200 volts/meter. With the 150fi/150pF of the standard ESD simulator, the field in the near region is predominantly magnetic. Because of this, the orientation of the victim circuit versus the potential ESD areas, and its source/load impedances arc determinant: A typical logic circuit wilt consist of a “victim” end being a logic gate input with input impedances of few kfi (TTL, Schottky) or more (CMOS), while the far end will be a logic output with source impedance ranging from 3013/15OQ (TTL, Schottky) to 3OOQ (CMOS). This arrangement makes the magnetic contribution worse. Decoupling the signal line at victim’s end wilt always be beneficial and matching the termination at receptor end has the same advantage. Otherwise, decoupling anywhere on the circuit may have no result, or even (I detrimentul one. Also, a fcrrite bead, though providing series insertion toss, may be disappointing because the bead should exhibit more added resistance than the gate input resistance, which is impossible to achieve with small size beads. To the contrary, if this is the power supply bus which has to be protected from ESD induced spikes, ferrite beads can be very efficient since the impedances are tow on both sides. A future study wilt investigate more deeply some aspects of the ESD re-radiation through seams and the induced voltages on ribbon cables running along those seams. References 1. Michael King: CORNELL Frequency I” MHz 1kll A ESD on protruding screw head, PCB terminated lilt0 B ESD on protrudlng strew into a short C ESD on vertical or horizontal seam with a forced PCB being 5 cm behlnd discharge point Figure 6-Voltage in an aluminum induced on the head, PCB terminated 10 cm x and Conclusions Finally, it is important that designers pay attention to the integrity of the metal housings, even for equipment which are neither RF equipment or highly sophisticated gear, but simply have to be ESD immune. 100 N 34 1mmgap, IO cm PCB trace housed 2. 3. 4. rack. 5. 6. DUBILIER Report on EM1 Susceptibility (December 1973). Peter Richman (Keytek): ESD Protection/Test Handbook. Michel Aguet: Perturbations dues aux decharges statiques. (COMPATIBILITE ELECTROMAGNETIQUE Ecole Polytechnique de Lausanne-1983). Michel Mardiguian: DESIGN FOR ESD IMMUNITY RATHER THAN RETROFIT--IEEE/EMC Symposium, Washington 1983. Donald R.J. White: EM1 Control METHODOLOGY and PROCEDURES (DWCI, Gainesville, VA 22065 USA). Michael King: Impulse Waveforms from ESD-IEEE/EMC Symposium. September 1982. - 35 7B3 - ESD Susceptibilityand Radiated Emissions of EDP Peripheral Printers Luciano Honeywell 201000 The E. S. D. emission the most stems We susceptibility (R. E.) and particularly ply with standard ship different between E.S. the relation- if described: the spark of mechanical gap sam- interferences electrically between floating electronic parts, (Printed differences only apparently xample, the E.S. a printer is then Last item GND bonding radiated D. and Board). mechani- grounded,are then such is outlined. As susceptibility level (inside the influence the printers) from EDP TEST Real-world, E.S. ted out in ref. H- field of effect pulse That’s and voltage is, drops, respectively, depend disrupted shielded peripheral the following D.air five routes: generated, ge induction H-field, from ground mechanical struc- effect, discharge current as a matter in two differents on E.U.T. as poinvia injection. of fact, items: predischar3. E-field, The last can be se5a. E-field gap, I. ESD loop; to only experience influence ex- discharge plate, internal enhance and the effects we’ve evaluation on nomena. For both lowing effect IEC laboratory l., and in the bus our 5a, methods 64 standard) and field diagnostic first on signal focused oriented, the printers voltage structure, discharge the out, in transient on mechanical Therefore, pointed on peripheral to be identified secondary 1)A effects: air generation generation spark as expeto enhance effect. main The of five direct or on E-field H. I, S. drops to the host by each gap for or on H-field has 1. predischarge, 2. spark methods, , can be used I. gene rated ternal on the ca- discharge in H. I.S. last of equipments radiation; E-field; 5. noise date discharge, disrupt different of METHODOLOGY (11, Two rienced As with emissions E. S. D. parated 1 current or strong. shielded computer ESD weak e- predicted. concerns connecting Circuit to understand phenomena on parts, between and a method troublesome the E.S.D. structural P. C. B. voltage analyzed, 4. tests. tire. The corona 5b. injected impedence ples; bles, 1‘ FIG. generators then, between coating .penerator. of E.U.T. effect, are, interferences unit of ES.! . L Test set&p for table-top9nounbd equlpmeht, field printer status :on IihC with compubr. test). items and surface cal Italy prin- and to com- D. structure under specific Pulse - many parts, no malfunction we analyse mechanical (equipment with regulations. first - Milan sy- E. D. P. environment In this paper The E.D.P. with to exibit and office - for concern be designed Two - now perhaps are topics peripheral devices, and electromechanical in home and the Milanese Italy peripherals. such ters; mechanical must Systems and radiated problems important Information Pregnana - Inzoli lines. analysis 5b, and 0 f ESD we defined two test with phc_ (folset-up, different purpose. one reference a common is characterised plane, GND star under point by: the E.U.T., for as the dischar- - ging system and E. U. T. 2) An electrical sible connection, between inside the plane the EUT, in ground data The power cord path of ESD The second 1) On line one The path power first then, cords set-up re- without (fig. of EUT, 1) by: frequency pulse, invol- purposes chanical equipotentiality structure 2) Voltage drops pressure are, printer level GND second star set-up systems surface (screws, of EUT as function purpose of are as following: l)Noise conversion unbalanced tive signal cable their transfer “Direct gap”: return to avoid its flat ground impedence “direct external spark discharge” H-f;eldESD and EUT cable .) effect in the case (n. 5b) and ca- and to arranged with mo re, “S”; internal Division a as shown C=150pF) oriented current spark with injection” gap “Sl”. it can be a T. D. R. Reflectometer, ARC stributed (R= lOOn, gap Further discharge, or lumped (Ti- R=50R ), to measure electrical The mechanical side only, has rent ways: Ni-coating;ZnCr04, of sample, been coating di- parame- grounded coated one in two diffechromate on electro mechanical and arc -deposited parts es can where and current ESD generator c ircuit V The equ iva lent with: delay-lines wave reference floating tact as parallel output I’ are inj%%d model guide finger and sample, to-end capa city. by IESD representative sample representative of the sample, res ist ence, excited current between CP re- voltage can be built-up (DL), plane, side ce coating be verified. is behaving circuit, E.M. can be evalua- discharge-surfa sample sonant T. “Interna&ark method, a flat impedence; discharge” oriented inte rferenc on or unshielded etc.. emission cable, generator can be: ted, of e- generated connections, can be: shielded 1. ESD printer struc- between radiated has 2. path, characteristic meters The (shielded oriented, in fig. as possible, cable, the sample Zinc (yellow and colourless conversion). In such a way, the oscillation modes of capaci- busses or fault connections comzison prevalent __I__ (i.e. mechanical or ground noise computer bles, to dif- frequency circuits). 2) Interference host of current between and power lectronic common due to high path coupling tures from mode, shown much conversion ferential structural ters. point. diagnostic invol- Therefore current as two mecha- method laboratory as short, me coatings). se t-up ESD the EUT analyzed. of a significant without on bonding 3) Susceptibility of me- of EUT. contacts, be first test The between and the test verified, The and cables. diagnostic must in fig. with connections. current ved control as following: 1) High The of EUT, condition of ESD in the connections. cables structure been characterised signal 2) Return ving is interferences on the or of pulse. condition cables running external filter The nical behavior connector. current signal point identified is not involved running external as pos- and the GND on line cable turn 3)Self-test short alternatively connection external as 36 between C, of RSURF ESD of and con- generator T,~return cable FIG. Test end- 2 set-up: Mechanical sample surface in g. with coat- - The results 1. can be Surface tor sure is face impedence 2OOgr. No bution is tact pressure lb, open are effect is The pres- flat has strongly 4. and screw exposure to Na Cl discharge nerator As point 4b.If corrosion voltage measures ESD gen. of smoothed input impedence point on a mechanical ranging then, if Cp=@, that’s sonance). The generator input equivalent loons). No electrical resonator is pointed return (std. pulse is used, and sheilded the current pulse. the 5,contact current shaand- pressure The very dangerous oscillations are spark ESD due to ge- gap S, series to disrupt related to the cable method) is is in fig. are circuited waveguides as fig.1 injected 6;the “derivated” cy oscillations generator arranged the typical shown shape and the ESD are triangular and excited set- current high by CRF clas- frequeninfluences. re depends- wn to lo- along structures if E-field, the If = 10; ESD or if its E-field 3 compling inthe in the case effect to significative rne- out. FIG. excited a mechanical floating coatings, discharge dependent. one Sl. If the sic fo=55MHz to surface from rela- to surface short resistence, up indicated, parasytic (2: “Sl” in fig. coating TIZD as it can be seen is deplaced to is on discharge classic of a second “S” related This is used, dependent. V parameter point correlation 4c is “S” frequency first : fo= 150 MHz coustant resistence, the to discharge resonance; dumping and no dependencies thod from a lumped impedence, the discharge related to 0.4KV. (i. e. a A/4 that’s is specific depends parameters cal high the peak structures: 1. 2KV coupling if Cp=80pF, out, oscillation at the from frequency, from ge- neration pointed cable gen. pe is shown (1 OKV) is used. value is dependent.ESD is an arc is a tringular coating VESD, parts. no correlation return shape test. 2. the and if ESD 04. at min., This is sample, as dis charge method IESD’ generator ( 10KV) is used. 4a.If holes. colourless, R,,,__=1;2Ofi(ty2;. 8fi&%% points). pically), Rsupr=* Ni: RSURF= + . 1; RSURF=. 05;. 5flafter 16h, point of mechanical coating. as RsURF=10;400~, edges parts. ESD is ranging discharge it is on flat is, dependent, is used. of oscillation ted phenomena; con- if coating, value on edge 0. 5KV contri- sharpness ( IOKV) 1. 2KV, fig.4,is exactly resi- edge peak from used. near 7B3 3 the sur reflections RSURF coloured, circuits lb. Zn Cr skin dependent, 04, contact a contact wave out. genera- in fig. merely to travelling pointed Zn Cr . shown significant been la and lc As - as 3* generator ‘ESD as following: TDR R SURF’ and a typical is needed, stence. la. summarized resistence 31 part (N. mechanical _-of prevalent 5-a) of a printer structure grounding’impedence (high frequency) of ESD effect is is rised values, the is the most FIG. 5 TDR 1 OOA/div pulse 20 ns /div. FIG. 4 Peak value ZOOV/div. 10 ns/div. FIG. ESD 6 current 1 OOA/div 50 ns /div. 38 This is a common coating with internal or grounding screws To test tion, are printed wire ference 1) with (as one mechanical capacitive than 1 pF. The results representative faced to printed is very surface input sharp mV). 1 limitations, Th: logitudinal the total V A VG zvIN is neglibible, more than ESD input be very &V will The to ESD is a real complex analysis matrix mechanical methodology has printer, with structure, been set-up a segment The H.I.S.I. significant parts, the following be measured 2a. electrical different with Characteristics tion times, delling such /REF. resonancies, must electrical a T. D. R. impedences of the equivalent bet- be pointed parameters As must lines trum mo- how resonators. GND. accurately pointed ILRNiZ repreand as noise can be easely is solved, on all by E.S. D. in Ref.[Z, value is given e- integration the spectrum. of the spec_ . The 31 by asymin dB, for each of time peak on line derived. value by the integration, in above in the pulse, program final of the referenced real will of phase one. the range information Many excess value redu- algorithm, in the significant error of tranbe a ve- papers,we or of the integration representation. mated system, current as approximated then due to the loss and propaga- pulse is shown approximation shown in the spectral a simple its ced the excess’err set-up: delay is obtained ry good out. the and the equation predicted sient, ween 2. All g scale in drops involved modulus log-lo an be deribed. domain for segments; transform the broad-band integration, ptotic set- 6. Its spec- in fig. circuits, ranges spectrum method be test approxima- Fourier of voltage P.C.B. A flow-chart, be or the preof 2a impedences. provided frequency will shown from connecting value ones, must domain voltage in the frequency Y vESD time of the injected for of 2 if are current, as can then peak choice by fig. if laboratory is be calculated source by mea- The by comparison pulse pulse significant approximated t zo) suscptibility susceptibility T. investigation 1. All representation, trum E. S. D. relatively of the tral sharp. poor! If the E.U. tion stimated, zo/(50 ESD a good of the inductance 2. results. up is used, sentation suggested step the prevalent indicated parts, connected”. parameters Typically, can band- value. = 2vESD. are inje cted devices suggested measured. 4. All or current no significative will 20, resonator The voltage and the printer very large. with (L is as ones,is the fi- noise is floating, can be very IESD, model, valent resona- high are of the 7 if distributed 3. The if different mechanical range values results diffe- devices. step by different parameter modulus, impedence part important by fig. Zn is cting or electromechanical and 2c measurement Only the can be very (AVClOO (Zo= and capaci- tonne electromechanical lurrped parameters of the R of Ni or Though impedence discharged guide and resi- inductances lines, “low-frequency surement less characte- (BW- Thisprotection is required against an
interferenzesignal whose freguency is stable
to within - 1 Hz of the tone spectral line
in the ILS tone filters.
The second level of protection ratio is 23 dB,
which is required when the interferencefreguency is stable to within - 50 Hz. In this
case the interferencemechanism results from
the action of the interferencesignal in the
,tonefilter rectifyingcircuitry.
The third level of protection ratio is 6 d0,
which is required when the int rference
freguency is stable to within 7
- 12 kHz. In
this case the interferenceeffect results
from the action of the interferingsignal on
the IF detectors.
,rherange of 40 dB between the protection
ratios quoted indicates that the protection
required by the ILS system is very dependent
upon the characteristicsof the interfering
signal.
!Xi/59
60/69Xl/79W/8990/99
Lf3.d at 30 m dB(uv/m)
Fig. 1 :
Histogram of measured spurii
levels from induction heating
equipment.
5.2 Characteristicsof ISM Interference.
The fourth harmonic of 27 MHz dielectric
heating equipment falls within the ILS band.
The characteristicsof the spurious signals
radiated by equipment were determined by
visitingfourindustrial sites and making
measurementson 42 separate dielectric
heating equipments. 'Theequipments split
into two main categories; those which operate
continuously (used in processes such as
baking and glue drying), and those which
operate intermittently(used in plastic
welding). The detailed results have been
reported C131.
From Figure 1 a level of 90 dB(uV/m) at a
measurementdistance of 30 m from the source
is exceeded in 20% of the readings.
5.2.1 Equipment which operates continuously.
Almost 50% of the equipments upon which
measurementswere made were of the type which
operate continuously. The actual fundamental
frequency used is adjusted to maximise the
transfer oE power into the load. The fundamental frequenciesfell within the range
26.06 - 29.88 MHz. 'Thestability+ofthe
fundamentalfrequency yas within - 10 kHz,
short-term and within - 20 kHz over a
five day period.
It is concluded that a satisfactorylimit
for the levels of spurious radiation generated
An assessment 'was made of the fourth harmonic
of the radiation polar diagrams of the dielec-
Many Regulatory Authoritieshave already
accepted CISPR Recommendationsconcerning
interferencefor equipment other than ISM
in which the limits are implementedon the
basis that up to 20% of the equipments in
use may exceed the stated limit.
44~2
237 -
tric heating equipments, four measurementsat
a given distance being made in orthogonal
directions. At a 30 m measurementdistance
the Iminimumvariation in a polar diagram was
10 df3and the maximum variation 35 dB. The
typical variation was 25 dB. At loo metre
Imeasurement
distance the minimum variation
in the polar diagram was 6 dB and the maximum
variation 35 dB. The typical variation was
15/20 dB. 'Theconsiderablevariation between
the measured polar diagrams of similar elguipments indicates the strong influence of
surroundingstructures on the radiation
influenceis not present in
pattern. ,I'his
test-sitemeasurements.
'Thelevels oE the measured interference Eield
strength at 30 m varied between 53 and 100
dB(uV/m). At 100 m the measured levels igere
between 27 and 69 dB(uV/m).
5.2.2 i___.____~-.
Bquioment &i-h operates
--i.ntcr,ni.tte!ntl.y
The plazic welding equipment radiated
spurious signals intermittentlywith
typically a 5 second "on" time and a 20/30
second "oEE'*time (while new material was
being inserted). The nower of the eguipment
was in the range 3 - 10 kW. The fundamental
frequenciesspanned the range 25.9 - 30.0 MHZ,
plus one machine operating on 20 MHz. The
EundamentaL frequencywas not stable, the
drift rate varying 'between11 kHz/sec to
200 kHz/sec. rypically the drift rates fell
within the range 50 - 75 kUz/sec.,
The radiation polar diagrams at the 4th
harmonic were estimated by making four
measurementsat a distance oE 30 metres in
orthogonal directions. The minimum
variation in a polar diagram was 10 dB and
the maximum variation 48 dB, typically
being within the range 20 - 35 dR. At the
30 metre distance the measured field
strength varied between 23 dB(uV/m),and
76 dB(uV/m).
5.3 Interactionof ISM Interferencewith
ILS receiver.
Some initial experiments in which typical
ISM interferencewas simulated suggested
that the ILS receiver could operate satisfactorily in interference Eields up to
20 d0 higher than the wanted field.
Therefore the interferenceeffects of the
spurious signals generated by actual
dielectric heating equipment on the operation
o.Ean ILS/VOR system was evaluated at two
typical industrial locations. One location
had 12 continuouslyoperating dielectric
heating equipments while at the other
location there were 13 bridge-typeplastic
welding equipments. 'Theexperimental
arrangement is shown in Figure 2.
The monitoring equipment was used to identify
the harmonic signals generated by the dielectric heating equipment within the ILS/VOR
frequency band. The amplitude and frequency
of the signals was measured with the
calibrated antenna connected to the spectrum
analyser. The calibrated antenna was then
reconnected to the ILS/VOR equipment and the
appropriatechannel selected.
Phe results obtained cl41 showed that the
protection ratio required for interference
oroduced by continuouslyoperating dielectric
heating equinment is of the order OF 2 - 4 dB.
'EhereEore,Eor such eqipment the protection
ratio of 6 dB mentioned in Section 5.1.should
afford complete protection. The results for
intermittentlyoperating equipment indicated
that t'neinterferenceeffects are present
Eor a very short time, sign.iEicantly
less than
0.5 sets. %'hereEoreit is diEEicult to assign
dn adeguate protection ratio but certainly
the nrotection ratio is significantlysmaller
than that Eor continuouslyoperating
equipment.
5.4 InterferenceLimit
During the crl'ticaltime when an aircraft is
just about to land the specified minimum
field strength for the ILS is 46 dBiB(uV/m).
Using the protection ratio oE 6 dR, and
making an allowance of attenuation through
the Eactory wall or roof of 10 dB, the limit
for TSM interference,measured at 30 m, is
50 dB(uV/m). In the derivation OE this
limit account has been taken of:the characteristics of the ISM interferenceand of the
response of the ILS system, but no account
has been taken of other factors such as the
probabilityof ISM eguinment being close to
the airport, the frequencygenerated being
within the passband of the ILS receiver or of
the equipment being operated during the
critical time. 'CheILS is a critical service
and therefore it is wiser to rquire that
I?(I)= 0. 'PIUSthe limit OE 50 dB(uV/m) at
30 m will.give complete protection to any
aircraft flying to within 30 m of a dielectric
heating equipment.
A2
cl
S/A
Al calibrated
PU
dipole
S/A
A2 monitor whip
SG
A
O-60 dB atten- ILS
uator
S
6 dB splitter ILS RX
DC power unit
spectrum analyser
signal generator
ILS/VOR tone
generator.
ILS/VOR receiver
Fig 2: Test setup to evaluate interference
to ILS/VOR receivers.
- 238 -
6.
CONCLUSIONS
The work of this paper illustrates two of the
important areas in determining radio interference limits. It is necessary to find a
value of the protection ratio required by
the radio service for the type of interference
considered. The other is estimating the
probability of interferenceoccurring iE a
particular limit is implemented. In addition
the importance of knowing the history of
interferencecomplaints in determining
suitable limits is shown.
Similar work is continuingwithin CSSPR,
most notably with respect to Data Processing
Eguiplnent.
It is hoped that by combining this type of
theoretical approach with practical experience,
the reassessmentof the ISM limits being undertaken by CISPR, in conjunctionwith CCIR,
will gain recognition!and result in the
limits being applied in practice.
I.
8.
REFERENCES
Cl1
CISPR Publication 11, 1975 and
amendment No 1, 1976.
c23
CCIR IWP l/4 - Doe. l/186 (Rev.11 - E
of 23 October 1981, Progress Report by
Chairman with technical annexes
5 - 18.
c31
CISPR/B
r-41
CCIR RQt. 829, 1982.
c51
Radio Regulation No. 1.60.
[61
CISPR/B/WGl (Whitehouse/UK)8,
February 1983.
c71
Radio Regulation 2857.
C81
Radio Regulation 2854.
r.91
CAP 208 Vol.1 pt.5.
Cl01
CCIR Rpt. 928.
II111
Dept. Trade & Industry DRr ,Yechnical
Memoranda Nos. 122 to I28.
L121
Royal Aircraft Establishment.
cl.31
ERA Technology Ltd Rpt.3572/5,
June 1982.
[14]
CISPR/B/WGl (Whitehouse/UK)9,May 1983.
(Secretariat)35, May 1984.
ACKNOWLEDGEMENTS
I wish to thank the Director of Radio
'Technology
for permission to present this
paper.
-
239
45
-
H3
STATISTICAL EVALUATION OF THE EMC SAFETY MARGIN AT SYSTEM LEVEL
BY
B. AUDONE, R. CAZZOLA and G. BARALE
AERITALIA - Avionic Systems and Equipments Group
Caselle T.se - TORINO - ITALY
ABSTRACT: EMC equipment tests are well es@
blished and defined, while EMC system tests
are still rather vague: in MIL-E-6051D dea
ling with EMC system testing only general
guidelines are given with the definition
of the safety margin. In a complex system
such as an aircraft, the problem of the
electromagnetic compatibility and safety
margin have become more important especially
with the introduction of electronic equip
ments into areas of the aircraft which
directly relate to flight safety. Interfe
rence effects are not always repetitive and
in many cases the malfunctions change ran
domly around an average level. Therefore
a statistical evaluation of a safety margin
was developed for the EMC investigation tests
so that a number of parameters, from various
avionic equipments could be monitored, when
the aircraft onboard emissive equipments
are activated. By use of this technique each
equipment parameter may be monitored over
a set period of time. Comparison can the
refore be made, from this point of view,
between each parameter with and without
the emissive equipments activated in order
to evaluate any drifting of any parameter
due to EM1 and obtain an EMC safety margin.
1. INTRODUCTION
In many works the need of a statistical
model for EMC testing has been emphasized
111, 121, 131, because the interference
effects are not always repetitive, but in
many cases the malfunctions change randomly,
around an average level. In a complex system
as an aircraft, the problem of the definition
of an EMC safety margin related to a stati
stical model is very important.
At the present EMC system tests are still
rather vague and only general guidelines
are given about the definition of a safety
margin. The purpose of this work is to
develope a mathematical model to perform
an analysis of random interference effects
from the statistical point of view and to
define a safety margin related to the chara:
teristics of the equipment under test.
Some considerations about the degree of
uncertainty related to the safety margin
evaluation are also given in order to reduce
the error probability during the execution
of the test at system level.
2. SAW'LE VALUE AND PARAMETER ESTIMATION
The two basic parameters of a random
variable x which specify its central tendency
and dispersion are the mean value and the
variance u and dL:
X
/X
(1)
-co
where p(x) is the probability density
function of the variable x.
An exact knowledge of the p (x) function
will not generally be available. Hence
one must be content with estimates of the
mean value and variance based upon a finite
number N of observed values:
(3)
The hats (-) indicate that
3 and axL
are
/.
used as estimators for the mean value and
variance of the random variable x. In order
to have a good estimator, it must be:
a)
A
unbiased
EC63
=9
(5)
-
240
-
n
where #
is an estimator of @.
b)
efficient
Qlx
2
1 is the Chi-square distribution function
n
with n=N-1 degrees of freedom.Moreover the
sampling distribution of the sample mean value
x is given by 141 :
n
where fl is the estimator of interest and
is &y other possible estimator of @.
@I
where :
- r
c)
consistent
where N is the number of observed values
and 3 is the estimator of $4.
It is desirable that the expected value of
the estimator be equal to the parameter
being established (estimator unbiased) and
also that the mean square error of the
estimator be smaller than for other possible
estimator (estimator efficient). Moreover
it is desirable that the estimator approach
the parameter being estimated with a proba
bility approaching unity as the-sample size
becomes large. The estimators u and 2 2
/x
X
are estimators unbiased, efficient and consi
stent for the mean value and variance of a
random variable x 14).
3. SAMPLING DISTRIBUTIONS
Consider a random variable x with a probabi
lity distribution function p (x).
Let x 1, x2, ......... ...... ..... .... XN be a
sample of N observed values of x. Any quan
tity computed from these sample values will
also be a random variable.For example,c3nsi
der the mean value x and the variance S of
the sample.If a series of different samples
of size N were selected from th? same random
variable x,the value of x and S computed
from each sample would generally be different.
Hence x and S are also the random variables
with a proba
P(x) and Q(S ).These functions are alled
2
"Sampling distributions" of x and S .
If the variable x is normally distributsd
with a mean of u and a variance of us' ,
the sampli?g diitsibution of the sampleX
variance S is given by 141 :
where :
-T
is the Gamma function
is the
Gamma
function
t = F(X-,uxJ/S
nn= N-l
Pit ( is the Student distribution function
wi& n=N-1 degrees of freedom.
4.
CONFIDENCE INTERVALS
The use of sample values as estimators for
parameters of a random variable has been
discussed previously.However those procedures
result only in point estimates for a parame
ter of interest.8 more meaninful procedure
for estimating parameters of random variables
involves the estimation of an interval,as 02
posed to a single point value,which will i"
elude the parameter being estimated with a
known degree of uncertainty.Such an interval
can be established if the sampling distribu
tion of the estimator in questi n is known
9
Y
based upon
For the case of the2variance d
a sample variance S computed fcom a sample
of size N,a confidence interval can be esta
blished as follows :
where:
r I
%;d\t
The degree of trust associated with the
confidence statement is 4-d
and it is
called "confidence coefficient". Furthermore,
if d 2 is unknown, a confidence interval
can s&l
be established for the mean value
ux based upon the sample values x and S as
/
follows:
-
241
45
-
H3
interference generators (emissive equipment
switched ON).When no interference occurs at
< 0) one may consider
system level (IM
the effect of thsYgmissive equipment (step c).
This measurement will still be referred to
the confidence interval ILo,Uol and ILo',Uo')
defined in step a) (see Fig. 1).
Therefore an effective interference margin
can be defined as follows :
where:
n=N-1
“Cm:dl2
The degrke of trust associated with the
and it is
confidence statement is l- d
called "confidence coefficient".
= 2010glo(Lo/L2) , for%iN,f;ji6(13a)
IM
EFF
5. TEST PROCEDURE AND SAFETY MARGIN EVALUATION
The procedure for estimating the interference
margin related to a parameter of interest has
been subdivided into three steps:
a) evaluation of sample mean value, sample
variance and confidence intervals for the
parameter of interest by means of the
specification characteristics of the equip
ment under test
b) evaluation of sample mean value, sample
variance and confidence intervals for the
parameter of interest with minimum number
of equipments switched on and interference
generators switched off (minimum noise co"
dition)
cl repetition of step b) with the same
number of equipments switched on and
interference generators switched on, one
at a time on the aircraft.
The step a) allow to determine the cop
fidence intervals /Lo, Uol from the
specification characteristics of the equip
ment under test. By means of the test
procedure recorded in the step b) one may
verify that the sample mean value and the
variance, with minimum noise condition,
falls into the confidence interval est.2
blished in step a). Fig. lA, B shows a
comparison between the confidence interval
ILo, Uol and ILo', Uo'I related to the
specification limits of the equipment
under test and the confidence interval
ILl, U11 and ILl’,Ul’I
obtained, for the
same parameter, with minimum noise con
dition. An interference margin at system
level may be defined as follows:
IM
= 2010glo(LO/L1), for XNC ZO
SYS
(12a)
= 2olog10(LO'/L11), for6:6;
IM
SYS
(12b)
IM
= 2010glo(u1/uo),
SYS
for Qx,
(12c)
IM
= 2010glo(U1'/UO'), forCi,$
SYS
(12d)
= 2ologlo(Lo'/L2'),forc+,<6+13b)
IM
EFF
h11.3O
= 2010glo(U2/Uo) , for%,;<(l3c)
IM
EFF
IM
= 2010g10(U2'/Uo'),for~$&.3d)
EFF
0
An interferent situstion occurs when IME& 0.
In this case [~2,U21 and (~2',U2'1 are
the confidence intervals computed with emis
sive equipment switched on.Moreover L. and
U. are the lower and upper limits of the
c&fidence interval for the sample mean
distribution function,while L! and U! are
the lower and upper limits of'the coifidence
interval for the
sample variance.
I
I
I
I
I
Fig. 1 A
I
I
I
Example of curves sampling distribution
function : (1) not interferent case(L *L
1,2 0
and U ._GU
) , (2) interferent case
1,z
CL1 *‘<
t
0
Lo)
Fig. 1 B
%'P4*,u;ri+,
I’
I'
I
I
Example of curves sampling distribution
function : (1) not interferent case(L' B L'
1,2
0
and U' 4 U' ) , (2) interferent case
1,2
0
(L; 2L L' )
0
An interference situation (IMSys>
0) in the
minimum noise condition is usually a clear
symptom of an integration malfunction to be
solved at system level. It is necessary
to remove the cause of interference at system
level before considering the effect of the
6.
PRACTICAL IMPLEMJXNTATIONOF THE THEORY
The evaluation of this confidence interval
in easily performed for a fixed level signal
whose level and accuracy are shown. This si
gnal may be represented as follows:
-
where :
K = nominal signal level
AK = maximum deviation
IM
All the signal values are included in the ra;
ge IK - AK ; K + AKI.
This range will be assumed as the confidence
interval of the sample mean distribution ,
with a confidence coefficient 1-d equal to
unity and a sample mean value equal to K.
ILo,Uol
IK-AK;K+AKI
=
-
= 201og ";
IM
loSTAT
L'
2
(14)
a(t) = K + AK
242
(15)
The maximum variance related to the signal
of interest may be evaluated considering
the following choice of signal samples:
S1 = K
S2 = K+AK
S3 = K-hK
Therefore:
STAT
= 201og
10
= 201og 1o
IM
STAT
,6;++$
u2
Ul
, x
U2'
Ul'
t 62,+,2CN2(18d)
ILo’,
UO’(
=
10;
4~~1
(17)
and the confidence coefficient 1-d is equal
to unity.
When the equipment specification does not
allow to obtain the mean value, variance
and confidence interval performed in the
previous paragraph, the measurement procedure
can be carried out referring to the confi
dence interval calculated with minimum noise
condition.
This is a worst case situation because the
confidence intervals [Ll, U11 and ILlI, Ul' I
should be usually narrower (IMSYsI fl) than
the confidence ontervals [Lo', Uo' 1 and
ILO’, Uo'I determined by means of the
equipment specification. In this case one may
define a statistical interference margin,as
follows :
L
= 201og
1
IM
STAT
10-,?N+*"
.L
2
2Irl
(18a)
(18~)
(19)
High probability of an interference exists
when IM
STAT' *'
The test procedure summarized previously requi
res the execution of a large number of measu
rements. In case it is necessary to reduce
the measurements number,only one mean value
(or variance) measurement can be made during
the test. The effective interference margin
can be defined as follows :
,;
,z
0
The minimum variance value is related to the
case in which all the samples are equal to K.
Therefore the minimum variance value will be
equal to zers and the mean variance will be
equal to AK /2. Therefore the confidence
interval of the sample variance related to
the specification limits is as follows:
N
= IM
- IM
IM
EFF
STAT
SYS
E
IM
= 201og
N+I
EFF
10U
N=3
ax
N+I
The statistical interference margin is rela
ted to the effective interference margin by
means of the following relationship:
L
IM
= 201og
EFF
lo+-x
N+I
where:
(18b)
IM
= 201og
10 &?EFF
N+I
4;;
o
ax
N+I
o
(20a)
(2Ob)
2
(2Oc)
‘&+&,
O?N+l
IM
= 201og lo ___,
Cr2N+I
EFF
Uo'
~~N+I';~(20d)
In this case the statistical distribution
of the values x
02N+I will be neglec
2r'I and
ted (see Fig.
Fig. 2
Example of execution time reduction
(1) not interferent case:* ore
I?+I 3
1
L
and x arc
I U 0rU'
N+I
0
0
2
(2) interferent case: x or flN+I ,( L
L 0rL
0
:,
0
or L'
0
-
243
The mean value
of a sample of N independent
/ux
observations of a random variable is assumed
as an estimator of the true mean value u .
Now the sample value u will not come bug
exactly equal to u 6"
ecause of the sampling
/O
variability associated with u .In order to
/
establish the probability of &is error,it is
necessary to specify some deviation of the true
from the assumed parameter u .
parameter
?O
/x
If the true mean value were in fact
=
+d
(21)
PO
Px
-
an error would occur with probability (3 ,if
the sample value u falls below the upper li
mit or above the ix
ower limit of the confidence
interval (see Fig. 3). By means of analytical
analysis 141,the probability (3 is related to
the samples number N,as follows :
S(tm;dlL
d,
where :
Ctfi;e)
'
(22)
I
S = standard deviation
n = N-l
1-d = confidence coefficient
d = maximum deviation
The previous considera ions are also valid when
the sample variance Q'
an esti
.x
mator of the true varifn:: assumed
:f120=df a + d; .It follows that :
n
o
the samples number N. Another error can occur
because the area related to the confidence in
terval is not equal to unity. The parameter
under test can assume values out of confidence
interval with probability d . Therefore dis
the probability of this second type of error.
The interference margin is computed with a un
certainty related to an error probability equal
to :
Prob IERRORI=Probl(TYPE 1 ERROR) U
(TYPE 2 ERROR)1 =
=0(+(-J-d?
45
-
in this section to emphasize the advantages of
the statistical analysis technique summarized
in the previous pages.In some aircraft there
is the possibility of using the onboard compu
ter to carry out the analysis previously de?
cribed. Each parameter is sampled with and wi
thout emissive equipment activated with the following input data :
- samples number : 500
- confidence coefficient : 95%
- time repetition or sampling :30 s
- data requested : x and S.
A performance safety margin typical of the pa
rameter under test is calculated according to
the expressions (20).Some examples of the test
results are reported in Fig.6 4,5,6.In this
case the system under test is the air data co2
puter (ADC);the emissive equipment is the UHF
transmitter. The ADC parameters controlled by
Test Integration Program (TIP) are :
- True air speed (TAS)
- Calibrated air speed (CALAS)
An interferent situation occurs (P.S.M.) 0)
when the UHF transmitter is activated. Thetest
has been performed as described previously in
order to reduce the test execution time. In
absence of specification data the P.S.M. has
been computed with reference to the confidence
interval evaluated with minimum noise condition
G,.
Fig . 4
(24)
Fig. 3
Diagram of the error probability area.
EXAMPLE OF THE STATISTICAL APPROACH
7.
Some experimental results shall be described
H3
Fig . 5
-
244
In some cases the analysis may be performed
by using a dedicated instrumentation. The di
gital signals on the data bus are analyzed
with a serial bus analyzer in the monitor con
figuration. All the data words transmitted
on the bus are checked and selectively cap$
red for the statistical analysis.
The serial bus analyzer can be programmed by
the user in order to perform a capture actor
ding to the flow-chart reported in Fig. 7.
Two factors have been used in order to select
a proper data word in the message :
- terminal address
- data word position in the message
Fig. 7
Data acquisition program flow-chart
For instance Fig. 8 shows a 32 data words
message transmitted by a remote terminal on
the basis of a predefined command word.
When the program runs on the bus analyzer
during the data bus activity,only the messa
ge corresponding to a set command word and
only a data word into this message,defined by
its message location,are captured. All the
captured data words are placed in the inter
nal bus analyzer RAM memory. At the end of
the capture phase all the samples in the RAM
memory are transferred to a host computer via
RS-232 or IEEE-488 for a statistical analysis,
according to the previous theory.
-
Fig. 9 shows a memory page in which it is pas
sible to see the captured data corresponding
to the sixth word in the messages of Fig. 8.
These data words are not equals for each sam
pling,but the values change sometimes.
Fig. 9
Example of memory page:the data words are
stored in the memory by inverting the most
and the less significant bytes.
This test procedure allows to discover all the
interferent situations produced by spurious
pick-up in the area before the encoding section
(external sensors,black boxes or remote termi
nals,interconnecting cables and so on) where
the signals are coded according to the 1553B
protocol and transmitted on the line. When the
interfere:iceeffect is directly related to the
on line signal(for example in case of impulsive
noise due to relays enclosure),this causes a
change on the 1553B signal waveform which results
in a transmission error. Also in this case a sta
tistical analysis of error generation can be per
formed by the bus analyzer by activating several
times the interference source. Fig. 10 emphasizes
a typical data transmission affected by errors
generation.
Fig. 10
Updating of terminals activity with tran
mission errors presence on the data bus.
Fig. 8
Example o~~~~~~~~~~F~s~~P~~:D~
a.
CONCLUSIONS
The test procedure described in the previous
sections may be performed monitoring some pa
rameters of interest one at a time. In many
cases the value v under examination is rela
ted to the other parameters
means of a functional relationship.For example:
is the
(25)
-
245
when 8 is interferred (IM
> 0), the
following situations are pE%ble:
- the interference is due to the transfer
function
dfl?f2
-.
.
) of the
equipment.
- the interference is due to one or more
parameters
In this
f$N.
case the equipment deos not.
produce further interference effects.
- the interference is due to the parameters
and to the equipment transfer function at
the same time.
Therefore in order to detect the cause of
an interferent situation it is necessary
to show the transfer function of the equip
ment under test and to test, with the emis
sive equipment activated, some or all the
parameters $ . . In many cases these are not
always accesstble for the test or the transfe _
function g ( di) is unknown. Therefore it
is necessary in this case to determine an
analysis technique which allows to discover
the actual interference source . This problem
has not been solved at the present and the
purpose on this section is to emphasize the
need to go back to the effective cause of a
malfunction, so that this will be avoided
easily.
45
-
-------------
H3
--
REFERENCES
Ill
C.W. Stuckey and J.C. Toler, "Stati
stical Determination of Electromagne
tic Compatibility", IEEE Trans. on EMC,
Vol. 9, pp. 27-34, September 1967.
121
D. Middleton, "Statistical-Physical
Models of Electromagnetic Interference",
IEEE Trans. on EMC, vol. 19, pp. 106-127,
August 1977.
131
H.P. Hsu, R.M. Storwick, D.C. Schlick
and G.L. Maxam, "Measured Amplitude
Distribution of Automotive Ignition
Noise", IEEE Trans. on EMC, vol. 16,
PP. 57-63, May 1974.
I41
J.S. Bendat, A.G. Piersol, "Random
Data: Analysis and Measurement Procedures",
Wiley-Interscience, 1971.
I51
MIL-STD-1553B, "Aircraft Internal Time
Division Command/Response Multiplex
Data BUS".
_. - - - - --
- - - - - ---
-
247
46
-
H4
THE STATE OF ART OF TV RECEIVER IMMUNITY AND RECOMMENDATIONS
FOR APPROPRIATE CONSTRUCTION, DEDUCED FROM TEST STATISTICS
R. Bersier
Swiss PTf, General Directorate
R 8 D Division
CH-3030 Berne, Switzerland
SUMMARY
The inanunities of 16 TV receivers of recent
design were tested by the "current injection"
method against disturbing AM and FM sources
in the HF and VHF range. From the results
good state of the art immunity levels are
established. The constructional features of
the tested receivers that could influence the
immunity are tabulated for the antenna input,
RF-tuner, IF and mains circuits. From these
findings reconunendations are made to improve
the construction.
1. INTRODUCTION
Recalling the problems concerning the RF immunity of TV receiver installations.
During the last years there has been a monotonously increasing number of complaints concerning the interfered TV reception, due to
the lack of immunity of the TV receivers or
of the installations. These interferences may
be divided into the following two principal
classes a) and b):
a) Interference due to the insufficient
TV receiver immunity in the short waves (SW)
range. The deterioration of the situation is
surely due to the increasing number of transmitters (amateurs and citizen band), that are
operating in residential areas. To this
interference class belongs the problem of the
increased licenced transmitter power and the
substitution of the amplitude modulation (AM)
by the frequency modulation (FM). In order to
clarify the situation, comparative tests were
modulation.
made
with
both
types
of
b) Interference due to the insufficient
immunity of the TV receivers or of the cable
TV distribution systems (CATV) in the reception channels.
It is well known that it is not possible
to reuse in medium or large CATV systems the
same channels that are occupied by local or
powerful regional TV transmitters: Due to the
bad immunity of the TV receivers and the CATV
itself, qhost pictures are created by direct
irradiation of the TV or the house distribution part of the CATV. A similar problem arises for the special TV channels (out of the
TV bands), if the transmitters of various
fixed or mobile services, operating in these
frequency bands, are placed in residential
areas.
up to now the problem was solved by the
appropriate choice of frequencies in the CATV,
i.e. free channels in the TV bands I and III
and non interfered special TV channels. But
the problem is becoming acute and difficult
to circumvent because of the constantly increasing number of TV programs to be transmitted and of radiocommunication services,
located in the interband frequencies of the
CATV (104-174 MHz and 230-293 MHz). Therefore
it is urgent to improve the innnunity of TV
receivers and CATV installations.
2. TEST METHOD
The immunity tests of the TV receivers were
made by the "synthetic" or "current injection" method, that is already described in
[I], [21, 131 and \41*
Recalling the principle of the current
injection method.
This method simulates the dominant effect of
the disturbing electromagnetic field on a
realistic installation, by injecting an asymmetrical current from a real source (Ri =
150 Q) through the connected cables to the
TV'S chassis. The immunity is specified by
the electromotive force of this source (E.m.f.
in dBpV) that creates a just perceptible
interference in the picture or in the sound.
Fig. 1 depicts the test set-up used. On
each cable (antenna and mains) of the TV receiver under test a coupling unit is inserted. The disturbing common mode current is
successively injected on each cable through a
source having 150 Q resistance, the other
cable being connected to the ground plane
through 150 8.
The relation between the E.m.f. value
that creates an interference by the current
injection method and the electromagnetic field
that creates the same interference in a real
installation was established experimentally,
~;~;o'l~;e'h~sso&l.inq
relations may be
:
1 to 40 MHz:
disturbing field(dBnV/m) 3 E.m.f.(dBpV) - 7dB
50 to 230 MHz:
disturbing field(dBpV/m) I E.m.f.(dBnV)
3.SCOPE OF THE TESTS
The immunity tests were made on 16 TV receivers with coaxial antenna input, system PAL
B/G. The receivers were of recent design
24e
receiver is 22 dB outside the TV bands, 34 dB
at the IF and 34 to 52 dB in the reception
channels.
b) The interference effects are more
pronounced with AM than with FM (difference
of 4 to 7 dB in the median values of the disturbance source E.m.f. that create a just
perceptible interference; the values refer to
the carriers according to sec. 3).
c) 50 % of the tested TV receivers may be
interfered by approximately the following
field strengths, when using AM:
115 dBnV/m (0.56 V/m) in the range 15-30 MHZ
109 dBnV/m (0.28 V/m) in the range 68-174 MHz
at the IF
81 dBnV/m (11 mV/m)
N 52 dBnV/m (0.4 mV/m) in the reception channels
(1981-82) and originated from different manufacturers, they are denoted in this paper by
the letters A to P.
The immunity was measured in the following frequency ranges:
a) 15- 30 MHz:range of interference from SW,
b) 68-174 MHz:range betweenTV-bands I and III,
c) 32- 40 MHZ: TV'S intermediate frequency(IF)
d) reception channels: 3 (54-61 ms),
S7 (146-153 MHz), 7 (188-195 MHz) and
12 (222-230 MHz).
In the first three frequency ranges, the
interferences produced by AM were compared to
those produced by FM. For this, the disturbing signal was modulated with approximately
1000 HZ in the following ways:
-by AM, at a depth of 80 % and then
- by FM, at a deviation of 5 kHz.
5. INSPECTION OF THE TV RECEIVERS' CONSTRUC-
Note: When applying FM, the interference pattern appears clearly more annoying with a
small frequency deviation than with a large
one.
The E.m.f. of the disturbing source was
measured using an average detector; therefore
its indication corresponds to the rms value
of the carrier and is not affected by the modulation.
For the tests in the frequency ranges a),
b) and c) the unwanted interferences that
could be created at the IF or in the reception
channels by the harmonics of the disturbing
source were prevented by the use of appropriate low pass filters or by the suitable
choice of the reception channels (see notes
in the diagrams).
4.
INSPECTION OF THE
IMMUNITY TEST
TION
OF
T3
Fig.
RESULTS
F
w : wanted
signal
signal
T2
Gl
Am
F
T2
Measurement sotup for
-1
Interference
RF signal
generator,
1.5-230
MHz.
Broadband power amplifier
(Sh: Shielded
box.)
Low-pass
filter.
Power attenuator,
6-10
dB, 50 fl.
thp
immunity
THE
IMPROVEMENT
r--z-;
3
Tl
----f-3cq~Gl
I
Metallic
ground plane
Mains coupling
unit
Rntenna coupling
unit
TV test
pattern
generator
FOR
IMMUNITY
5.1 Influence of the screening of the antenna
input circuit and of the tuner.
Pursueing the disturbing current,injected on
the screen of the coaxial antenna cable, we
encounter successively the antenna connector,
the optional insulating capacitors, the cable
to the tuner and finally the input of the tuner. Then the disturbing current is distributed on the screen of the tuner before arriving on the chassis ground circuit of the TV
receiver.
The value of the interference voltage
produced at the input of the tuner will be
directly proportional to the sum of the
transfer impedances of the components of the
input circuit; that is why these impedances
should be reduced to a minimum. The care taken
to the screening of the tuner (elimination of
slots, well contacting covers) and the performance of the various feed throughs (filtering) will be decisive in preventing the penetration of disturbing signals.
The above considerations are reflected in
the diagram 4, representing the immunity in
the reception channels:
i = interference
pJ-@-P
1:
RECOMMENDATIONS
Fc I
Pl
M
A
P
AND
THEIR
The constructional details that may influence
the immunity are indicated in table I. Comparing these to the diagrams 1 to 4, we make
following observations:
The E.m.f. values of the disturbing source
that created a just perceptible interference
are indicated in the diagrams 1 to 4 in a
statistical representation. The corresponding
values of the field strengths that would
create the same interference in a real installation are given on a parallel scale. These
values were obtained from the relation given
in sec. 2.
The inspection of the diagrams 1 to 4
shows:
a) There is a large spread in the results:
The difference between the best and worst TV
I
-
test
of
L-
- I
T
-_I
Sh
TV receivers by the current inj
and (Utij ).
Thus, in order to determine a conventional function of probability distributions Fd+ it is first necessary to find a composition of lognorma1 laws.
As shown in [I] , a determination
of a composition of these laws is
unfeasible in an analytical form because a characteristic function of
these laws is expressed by an infinite series. It is proposed in [I] to
use an approxinlatiorlof lognormal
laws by means of gamma functions
with the aim of finding of an approximate solution of this prob?.im.
In so doinn.
wheref([&
P
and (r+l)
are tl;anmla
functions;
are parar!letersof+a
probability distribution.
is aasily found in a
similar way.
In order to estimate the error of
an accepted approximation the values
of IJ,, were calculated, which cosrespond to comparatively high levels of
probabilities, F ~u,&o.o
- 0.95,
that are used fox es! imation of quality of radio communication. To this
effect, by a method of numerical integration on a computer, values were found of
g ability distribution
functio%R6° #Y mormal laws with
equal values of-standard deviations
and with K=2;4;8;16 and of corresponding probability distribution functions which were obtained as a result
of approximation,
The results of calculations showed
that within a wide rank;e of values of
standard deviations of effective interference voltagfes, produced by individual source~,W{U~i~~
6 - 15 dB
and that the maxirrml differences between the values of UQ+ccdo not exceed
1.0-1.5 dB.
In order to determine a probability distribution function F,,it is
,
47H5
- 257 -
necessary to find a compositior$ of
distributions F_C U&and
account the accepted
approximation the characteristic function or a probability distribution
Pti,K
is
whcrt: [p,,(1(;+4),aJ and Ifl&z*i),a21
are, respectively, the generalized
parameters of probability distributions of values of interference from
individual sources of group one and
group two at the input of a radio
receiving device. The values of the
parameters aredetermined depending
on values miandWL from graphs of
Pig.A.
Taking into account, firstly,Formula (r), secondly, the probability
distributions for the number of sources whose interferences act at the
input of a radio receiving device,
and thirdly, those probability distributions of the nurtlberof interfcrences belonging to group one and
group two, which were accepted in
Section 3, we shall obtain an expression for a probability distribution
function of effective voltages of
total processes of interference which
penetrate out of electricity supply
networks and through the field:
cpeq3’
when
K=d)
;
p and g are, respectively, the probability of appearance of
interferences of group one
and group two among the total
number of B interferences
which act at the input of a
radio receiving device.
In a particular case, when at the
input of a radio reoeiving device the
interferences of only one group act,
for instance, the interferences radiated by electrical equipment, the
formula is simplified:
5.
Permissible values of total interference processes at the input of a
radio receiving device can be determined from Formula (6) and condition
where F-'(d)
is suci a value of the
argument of a function
which is determined by
parameters which are
included in expression
(6);
In order to estimate a possibility
of determination of permissible values
of man-made radio interference from
individual sources let us use a simplified Formula (7) taking into account
condition (1) and a formula for b
which was given above.
when 0 < U2
ej& ka,pd+(N-K)aepg
In Formula (6):
cf
i
is the number of combinations of
N things K at a time;
is anjaverage number of interferences which act at the input of a
radio receiving device;
z = 7-+& - kca4&- Wic)a&g
,
In the CISPR, the limit$ o?interference are expressed in quasi-peak
values. 'Therelationships between
effective and quasi-peak values (acco.
rding to the CISPR Publication 16)
are in sufficient detail treated-in
work [4]
It is ihown there that the following conversion factor can be introduced:
Kr * 20 1%
.A& - 20 1%
Qyp
"i eff
-
Kr * 3 i 2 dB
for a frequency band
1 O-150 kHz;
Kr m 15 f 2 d% for a frequency band
0.15-30 MHz;
Kx - 1'7 2 2 dB for a frequency band
30-1000 MHZ.
When determining the permissible
values it is also necessary to take
account
- of the difference between the
bandwidths of the radio receiver
and of the xadio interference measuring set (A fz, and A f+,e,res-
258
-
6. Discussion of the
results obtained
The results obtained enable to
c alculatc the permissible values
of total man-made radio interference processes and to establish requirements for interference from individual appliances.
References
pectively),
- of the probability level at which
the limits of man-made radio interference are specified.
Taking this into account we shall
obtain fox quasi-peak values of interference:
CISPR/AWG2(Pevnitsky-USSR)2,
April 1978. On the draft CISPR
Report on Study Questions Nos.
54/l, 55 and 77. International
Electxotechnical commission,
International Special Committee
on Radio Interference (CISPR),
20 p.
Pig.2 sho& the results of calculations of Fq(d) which are included
in Formula (8).
AJl .KamaKoB
F;:)
?5
'~acnpexeneHm
Bf3pofiTHocTei2 Hanpmemi;
pamonoMex
Ha axoAe npuemmrca nprn owospeMeHHOM J303fie~cT~MM CJfyYaBHorC YMwa
~~~~MTHuX
CMrHeaOB". Tpy~qa
HlilVP
181
4) 19B3r.
:i.E.iiiop
C~wmcwdecwe
MeTow am
JIMSa M KOHTPOJI" ICaWCTBa M HQeEHOCTM.
~OH,PiZ~~O, ~:OCtiB+%?.
B.lI.rlemI4r~Kr4il"I{ sonpocy 0 COOTHOWf?HMJ3[X MelEJQ~ KBa3WlUlCOBHMM M
e$$eKT'GIBHHMM BHaYeHMRMM
KB%3MUMl-@bCHhrX I'IffOIJeCCOB
pazqvlonmex 0 Tpym
-
259
48
-
THE APPLICATION AND DEVELOPMf8NTOF Em
Q.
~6
IN CHINA
Chen
Y.C. Zhu
China AtiationResearchIn&.tute for Standardiaation
Beijing,China
This paper introduces comprehensively the application and development
70's to the early 80's, China possesed
of EMC, including the new efforts in
engineering development, measurement
teohnique,standards and SpeCifiCatiOnS,
EMC training in industrial, Scientific,
which had
medical, aeronautical and ship building
area in China, and EMC international cooperation.
ried out for 4 years to study and analyse the condition of electromagnetic
pollution and interference in China.
~&XXXHF plastic hot-jointing
caused a great
interference
for lack of perfect shielding, ground..
ing and strict spectrum OOntrOl.InVeStigation and measurement had been
In 1970'9,
electromagnetic
ference caused
General
machines
by
various
car-
inter-
radio-fre-
quency equipment used in industry,science and medicine interfered in
quirements for anti-conducted interfew
2596 of
coverage area of TV broadcasting of main
nce was placed on aeronautical industry,
large and middle cities, so that the TV
electronic industry, and ship building
etc. in China. But for radiated inter-
sets could not get clear picture.During
ference, the requirements for electromagnetic susceptibility didn't come to
of aeronautics, ship building,broadcasting and TV,labour and environment pro-
In 50's_60's of this centry, the re-
a decision to prepare standard then.
Since 1970's, the application and df+
velopment of EMC in aeronautical,
ship
this period the
departments concerned
tection, and railway transportation had
made a number of live measurements
and
statistical analysis of electromagnetic
building, television, communication,and
interferenoe, prepared or revised stan-
electric railway
dards, in which allowable
transportation
areas
have progressed rapidly. In this period,
limits
and
ference, the performance of some equip-
test methods were established. In the
same time, EMC engineering design and
measurement technique had further deve-
ment
loped. The quantity of researchers
because of intra- or inter-system inte*
degraded to such
they could not
an extent
that
work normally. Electro-
magnetic transmission with
large power
subjected the inflammable and explosive
materials to a very hazardous situation.
In order to eliminate oocured electro-
engineers who engage in EMC had increased greatly. A lot of progress had been
made in respect of test and measurement,
standard and specifioation,analysis and
prediction, research and development.
magnetic interference,great expenditure
and time must Sometimes be spended. For
example, according to statistics, from
and
Research and Development
-
260
In 1980's China had already applied
the EMC analysis and prediction techniques to aeronautics, telecommunication
and 90 on. When the multiplex system
was used in aircraft, it was found that
EMI caused by electronic and electric
equipment is the main cause for operation failure of this system. The way
in with multi-dimensional random process can be resolved with computer was
-
mated only by describing with quasi Peak,
when estimating interference characteriatics of analog and digital
system.
The MD
COtmmiCatiOn
measurement has been
adopted in China. The
develooed
measurement instrument in
APD
conjunction
with microprocessor have many functions
such as control. computation, sampling,
data processing, display, typing and
adjusting measurement time. The results
found, thus the code error rate caused
by conducted, radiated interference and
of measurement may be displayed by
spike can be predicted, providing thereby a scientific basis for reliability
design of the system.
in decimal number. It also can be plot-
Amplitude
or be typed out by microlattice hrpewiter
ted out by x-y plotter or recorded
tape or magnetic disc.
probability destribution
(At?D)measurement
interference is
of
electromagnetic
the most advanced mea-
surement method at
present. It is also
an effective way for
The structure
and
establish
has been
proved that effects of inter-
ference could not
It
be effectively esti-
Compar&tar
I
1 bAefOJ”nce
Amp
I
Vo/tayO
AI
IF
Fip i.
blocK d'lagram
APD instru-
input signal: intermediate frequency
output (-" 2.5MHz) or
peak detected output
measured item:It can measure Am and
NAD simultaneously to
10-4s for APD and to
IHz-40KHz for NAD
measuring level: inl2steps, sensitivity of the corn...
parator: 1.5mv
interference
mathematical statistics.
digram of AFED
The main performance of
ment are as the following:
the effects of interference on communication system. This method can be used
to find out the nature of interference
source
block
on
instrument is shown in Fig 1.
people to study
model by
LDE
of APD
measuring
instrument
- 261
measurement
times At IF of I.67 ~~IZP
the time for APD
measurement can
reach 1285 seconds;
at IF bandwidth of
12oKRz, up to140
seconds for NAD
48
-
f erence equipment, susceptibility test
signal source, and RMI/RFI dattiacquisition system, manufactured by companies
in West Germany, U.S.A. UK are
S,Scifications
and Standards
_-_)___~,_,I-,_____-_-~__"_-
BYD
circuit, including 137
digital
chips used as comparator.
With interference produced in Cities
and on electrified railway,ApD measurement have been successfully conducted.
The hazards of electromagnetic energy
such as electrostatic hazard, hazard to
inflammable and explosive materials and
effects on animals are subjects under
studies in labour and environment protection and medicine areas. The EMC techniques are adoped in hardware design
In the early 60's the standards
electric equipment, and standards on
their test method were released. SPecifications and standards have developed
rapidly, composing a complete series
of ErilCstandards and specifications.
Among them, the primary specifications
are
China has completly adopted the measuring instrument specifications prepared by CISPii,All publications No.l-16
of CISPR were translated into Chinese
RF anechonic
chamber of various typies and sizes with
conical or rectangular shapes were built.
Their working frequency ranges are laid
on P through X bands.These cell.3can be
used to perform simulating measurement
of air space, to study RMC of system,to
test the radome and to measure antenna
In the meantime many related
national
general
advanced
standards
ment,
the measurement is
conducted in
standards
inter-
and
world
have been studied in
detail.
For radio-frequency equipment
system and radar cross section etc.
With respect to El% of marine environ-
following:
tl3
Requirements for electromagnetic interference and electromagnetic
susceptibitity,
Measurement methods,
Definitions and System-of-unit,
Lightning protection,
Static charge protection,
Electromagnetic compatibility of
system ( EMCS ),
Specification3 for electromagnetic
interference measuring instruments,
EhlC requirements for electronic instruments,
Safety standards of microwave radiation.
Measurement Technique
A number of
on
the conducted radio interference characteristics of marine r?ndairborne instrument3 and radio apparatus, household
by machine building industry.
and issued,
also im-
ported.
The midium size chips are used thmughout this
~6
dustry,
science and medicine,
electrical
equipment,
radio
in in-
household
and
T.V.
the test field simulating the seas. The
receiver, mobile vehicle
Coupling measurement and the characteristics of radiation patterns are pre-
and high-voltage
dicted through simulating by scale down
model.
publication No.16 of CISPR are applied.
The TEI'II
cell is already used to measure E!{Cto study radiated interference
system,
power
the documents
and
igniter,
transmission
of II%:,CISPH end
For zhips,"The Proposal of Ship EMC"
is used.
This proposal delineates cri-
teria to be applied to verification of
EM1 and anti-interference. Test freq-
and charactristics of radiating source.
Nowever,this cell is mainly used to calibrate measuring equipments.
As f Or measuring equipments,in addi-
uency ranges from 0.01 to 1000 MHz.
For aeronautical equipment, the documents "Terminology of EMI and EMC"
tion to domestic products,
and
some equip-
ment, including electromagnetic
inter-
"EMC Requirements
for Aircraft
and Test
Equipment"
are
Method
applied.
- 262 -
These standards specify in detail the
emission limit and susceptibility of
conducted and radiated electromagnetic
interference and their test method. The
item to be tested are 13, including
conducted interference of power, signal
and control wires,and anttena end: conducted sensitivity of power wire and
control wire: mutual modulation, cross
modulation, and sharp peak signal: and
radiated interference and its sensitivity. The equipment are classified intO
categories as specified in these specifications, so that different items can
be selected to test according to category of the equipment. The requirements
for compatibility of system delimitate
a 6 dB safety factor of electromagnetic
interference, but for inflammable and
explosive materials this factor must be
20 dB.
China takes great interest in adopting international general standards
and world advanced standards. Some EMC
standards of IEC, CISYK, MIZ, ISO, HTCA
and BS etc. have been widely issued and
selected for use in purchase contracts.
the standard correctly.
The short terms lectures are jUSt
like lectures given in college. The
main contents .forthe lecture contain
introduction of EMC, hardware engineering design,standards and specification%
methods of measurement, analysis and
prediction ( software ). The Person8
attending each lecture counted 100 to
ZOO.
The coverage of academic discussion
ard wide, the form are different. Since
discussion has
1980, national academic
been held twice. Usually the meetings
of different industries are held more
frequently. People who attend this kind
of meetings are in hundreds. Local EMC
symposium are also held more freqently.
In these symposium, protection of EM1
caused by HF plastic hot-jointing machine, the interference of electric locomotive on receiver, mutual interference
of inboard equipment, susceptibility
EM1
measurement, and measurement of
safety factor were disscussed.
International Cooqaration
Since the late 70's, China has
EMC Training
already begun to exchage EMC technique
Over the years, a EMC tranining has
with many countries and international
been given to the people who are engaorganizations taking part many times in
ging in design, test,production control
activities
held by CISPE, ISO and IEC.
in various ways. The scope of training
The
chairman
and experts of CISPH and
includes basic theory, guideline8 for
SAE visited China and delivered lecture
engineering design, test method, stanand heldacad ozic discussionwith &he Chinese
dards, specifications and project plan
and control.EI% training has made those
experts. The Chinese experts and engiengineers understand and use EMC technineers are well informed of transactions
ques correctly at each phase of design,
of EMC symposium and exhibition and EMC
production, test and maintenance.
proceedings of IEEE, They are very
Methods of training are
interested in these information.
Et$g;Elish booklists and magazines
As described above, China has imported
several complete sets of EMI/EMC
to givi short terms lectures,
to test, demonstrate and exchange
equipment made by some campanies of the
views on the-spot,
U.S. UK and Westen Gemany etc. OF all
to give instructions in interpretaothers they are H/S carp, HP carp,
tion and application of certain stanE.T.N. COHP, SINGER Corp. The equipment
dard or specification,
to Solve problems of EM1 with EMC
imported include interference field
technique,
strength meter, spectrum analyzer,
to introduce application and popilari.verioua signal sources, detector, data
eation of new EMC technology abroad,
acquisitionand analysis
to hold EMC symposium.
system and
Before and after a new standard is
software pertinent to them. Their operissued,we usually hold meetings to diaation frequency is up to IH GHz. They
cuss some technical problems. At the
have been used in many departments in
meeting, the people who prepared the
China such as aeronautical and shipdra.fts of the standard shall explain
building industries, broadcasting and
and verify some
important problems by
T.V., traffic and tele-communication
necessary demonstration and calculation
etc.
to make persons concerned understand
-
263
49
-
I1
NEW WAYS FOR INTERFERENCE COMPUTATION AND MONTE-CARLO-OPTIMIZATION
TO GUARANTEE THE COMPATIBILITY
OF INDUCTIVELY
COUPLED
LINE
SYSTEMS
H.-J. Haubrich
Vereinigte Elektrizitdtswerke Westfalen AG
Dortmund, Fed. Rep. of Germany
Summary
The electromagnetic compatibility of the
components of power supply systems is a criterion participating more and more in the decisions of network planning and operation. In the
following problem cases of electromagnetic and
electrostatic induction by alternating fields
at power frequency are treated, caused by power lines of the electricity supply system to
lines of the own and of external energy or information transmission systems.
The basic idea is to transform the,inducing
voltage Up and current Ip of the field producing conductor P (e.g. the high voltage line)
into an injected current IpV with help'of the
mutual inductances and capacitances LPV and CpV
between the conductors P and V. Line V can be
an element of an extensive meshed network V consisting of pipe-lines, telecommunication lines or high voltage lines. Tnis network is modeled by its admittance matricYV, thus enabling
an easy representation of any optional topology.
With respect to a possible impairment of technical installations not belonging to the electrical power system, special attention has to
be payed to three-phase overhead lines as a source of electric and magnetic stray fields. They leave the closed electric substations and
meet with practically all other line systems
when crossing wide regions.
The principle of trace bundling followed by
the licensing authorities in Germany forces the
power lines to have long parallel runnings with
a corresponding strong coupling to other lines.
High voltage overhead lines are technically and
economically predestinated for high transport
capacity. Due to their high operating voltages
and currents and extraordinary high short circuit currents they can produce interferences in
quite a wide range.
@mpensator
Fig.1: Electrically coupled lines with given
inducing directions
The calculation scheme is independent of the
feeding conditions and of the number of disturbing lines P: the resulting source current is
obtained by geometric addition of the single
components.
The view of the acceptance or inadmissibility of these interferences requires the quantification of the physical effects. The analytic
simulation leads to reliable forecasts with respect to system stresses and suitable countermeasures. A universally valid algorithm for the
calculation of interference voltages and currents in networks of optional topology or with
several exposures to power lines is presented
and applied to practical problem cases. In combination with the Monte-Carlo method, one can
optimize the protective earthing of pipe-lines
against dangerous induction voltages.
I
A universal algorithm for
interference calculation
The capacitive and inductive interaction of
coupled lines according to fig. 1 can be calculated with the uniform algorithm shown in
fig. 2.
i_______+ Coupling
I
Matrix v-c
Fig.2: Computing scheme with injected currents
264
When several lines of the network V are simultaneously involved in an exposure, current
sources have to be added at the boundary nodes
of all interference sections. The resulting
network equation
-
Both above mentioned constraints can't be simultaneously be fulfilled within the permitted
mistuning range. There is a demand for additional provisions to guarantee the compatibility
of both voltage levels on the same tower, e.g.
by transposition of the 400-kV-circuits [3].
is well suited for the computer calculation of
the required nodal voltages WV to earth and the
induced branche currents IV.
A necessary presupposition however ist the
existence of homogeneous line sections with uniform exposure to the inducing power lines
which have to be approximated by sudivision in
case of need.
Additional extraneous earthed conductors C,
e.g. earth wires, metal cable sheaths or special compensation conductors, are often involved.
Their induced currents IC, determined in the
same way as IV, additionally act upon V by the
coupling C-V, diminishing the interference in
case of zero-sequence components IR, but partially also increasing in case of symmetrical operation of the disturbing three phase system P.
The reaction from V to C cannot be generally
neglected; in case of a close proximity V-C,
the dashed feedback in fig. 2 has to be regarded.
Point
Fig.3: Neutral point displacement voltage
Uo(Ir) of a llO-kV-network with capacitive coupling to 400 kV lines
Electrostatic induction in high voltage
networks with resonance earthing
Resonant earthed power systems must fulfill
two main restrictions with contrary demands for
the tuning of the arc suppression coils:
1.
in case of a single phase-to-earth fault,
the residual earth current Ir should still
ensure the self extinction of the arc (e.g.
I, < 130 A for the llO-kV-voltage-level cl]),
Electromagnetic induction in
pipe-line networks
Modeling
The mathematical model of a line with uniform exposure to the magnetic field of a power
line is derived from the line element ds in
fig. 4a.
2. under normal operating conditions, the neutral point displacement voltage U, should be
kept as small as possible (e.g. U, < 10 kV
in llO-kV-networks).
_,Fig.3 shows the calculated geometric locus
[2] of the voltage vector U, in an IlO-kV-network when mistuning the arc suppression coils
during normal system operation. U, results from
the capacitive induction by two 400-kV-circuits
installed on the same towers with two circuits
of the IlO-kV-network for a length of lpv=53 km
Resulting from the geometrical unsymmetry of
the phase configuration the 400-kV-circuits transfer a high zero sequence voltage into the
llO-kV-network even if the three-phase inducing system is balanced. The equivalent injected
currents are easily found, when the 6x6-matrix
Cpv of the mutual capacities CPV per unit length is known:
i,,
=
jw
lpv*
C,;
u,
(2)
Fig.4: Equivalent circuits of homogeneous lines
with inductive interference
a) line element ds with the induced
e.m.f. EpV
b) uniformly exposed long line with the
induced current source IpV
YV:propagation coefficient of line V
WV:surge impedance of line V
265
The induced electromotive force EPV per unit
length
E
PV
iEn
= c IPi'ZPVi
(3)
P
-
49
The exact line equations of a
posed section lv
coshyv 1v
LI
U
Vl
E
includes the contribution of the np inducing lines P. When calculating the mutual impedances ZPV between the lines P and V by
the Carsson [S] formulae even oblique exposures
may mostly be handled as equivalent parallel
lines in the distance Jal'a2 (fig. 5).
PV
uniformly ex-
WVasinhYVIV
uv2
.
=
Wv'IvPyv
I1
sinhyv 1v
w
WV'coshYVIV
I
L
v'
I+%
v2
y,
I
yield immediately the required injected currents
IPV
(5)
= EpV/(~V.WV)
representing the inductive interference in
fig. 4b. After dividing in line sections of
quasi uniform exposure any network can be composed by such basic quadripoles which define
the nodes and branches of the passive network
model. y,.
i
iwm
lil
middle
gmn.tric
dtrtmc.
lioo
w
-
Fig.5: Failure F of the e.m.f. induced in a conductor V with earth return if modelling
its oblique exposure to the disturbing
conductor P as paralleliSm(frequency:
50 Hz;conductivity of the earth: 50 R m)
Real example
The efficiency of the described algorithm
shall be demonstrated by the example of an underground pipe-line network whose three branches are exposed to the magnetic induction of a
400-kV-double-circuit line (fig.6).
L*>,‘\
\
/’
/’
1;=1,2kA=I,b
vv=(O,O22+j 0,047)/km
W,=h3,0
+j 2,4 ) R
Fig.6: Real example of a buried pipe-line exposed to the steady-state magnetic
field of a 400 kV overhead line. Construction of the sections with quasi
uniform induction.
0
: node number;
- 20 --:distance in meter
a
C
-
The length of exposure reaches more than 10 km.
Following the discontinuous line configuration
and coupling impedances along this length, the
pipe-line has to be divided into numerous uniform sections corresponding to the vertical propagation of the power frequency magnetic field. The dotted reference axes show the way how
to determine the nodes of the pipe-line model.
The induction by a single-phase-to-earth fault was proved to be less critical than by the
balanced loading of the 400-kV-circuits. That
induces pipe-line voltages to earth exceeding
by far the permitted value of 65 V (fig.7, curve a). Without suitable protection devices maintenance staff working on the pipe-line would
be exposed to danger.
266 -
of this so called Monte-Carlo method includes
the costs CF for the protective earthing at m
feasible ground connection points (fig. 8a) and
the constraints uVk< 65 V, the violation of
which is punished at all regarded nodes nV by a
penalty function PF according to fig. 8b.
CF
CF
\
L
---_-_---__-_
w
0
a)
conrtrrlnt
i.
0
R
65V
t11
“V
Fig.8: Examples of the cost function CF and of
the penalty function PF for the MonteCarlo optimization
UV 0 65 V
Start
Define
Fig.7: Calculated induction voltages line-toearth of the pipe-line act. to fig.6
I
Calculate OF Ir,) - two)
I
a) without any countermeasures
+ PF(u,)
-t
b) with an optimal earthing resistor of
35 51at node number 10
Chance variation of rg -
I
I
Calculate
Optimization of the protection devices
1
vector r.
starting
r
OF(r)
I
1
I
no
Protective earthing is usual and well suited
to damp the dangerous potential rise induced by
the steady state magnetic field. The search for
the most effective earthing points and earthing
resistances may be very complicated since the
low-resistance earthing of the normally well
isolated pipes jeopardizes the efficacy of the
cathodic corrosion protection.
no
Nowadays computer-aided methods allow an automatic optimization of the protective earthing
by lumped resistors. For the case in question,
the method of statistical trials has proved to
be useful [4]. The objective function
OF
= C CFi
+ C PFk
is m; k.s nV
(6)
Solution
CL3
r
Fig.9: The Monte-Carlo search process for the
optimal resistive earthing of induced
pipe-lines
-
267
CF and PF are functions of the earthing resistors Ri, being the control variables of the
optimization process. Fig. 9 marks the main
steps of the iterative Monte-Carlo solution. An
arbitrary chosen starting vector I,=(RI,...Ri,..
.Rm) delivers the reference value OF(r,), that
shall be improved by a following chance variation of all random variables Ri within a given
margin. A sufficiently optimal solution is reached when subsequent iteration steps don't
bring any more remarkable improvement. In numerous applications the convergence was reached
after about 20-n" experiments, bad trials inclosed.
The application on the above shown problem
case yields a 35-R-resistor connected to node
10 as the optimal protecting measure.
49
-
I1
References
[l] VDE 0228, Tail 2/7.75: VDE-Bestimmung ffir
MaRnahmen bei Beeinflussung von Fernmeldeanlagen durch Starkstromanlagen, Beeinflussung durch Drehstromanlagen
[2] Poll, J.: Sternpunktverlagerung in gelbschten llO-kV-Netzen. Elektrizitatswirtschaft
80 (1981) H. 22, S. 810-813
[3] Brandes, W.; Baubrich, H.-J.: Sternpunktverlagerung durch Mehrfachleitungen in erdschlui3kompensierten llO-kV-Netzen. Betrieblithe Erfahrungen und AbhilfemaRnahmen.
Elektrizitatswirtschaft 82 (1983), H.ll,
s. 400-405
This mathematical solution is at least a
helpful approximation, even if the implementation may be modified or adjusted by further practical considerations.
[4] Schwefel, H.P.: Numerische Optimierung von
Computer-Modellen mittels der Evolutionsstrategie (Basel,Stuttgart:Birkhauser 1977)
The high efficiency of the described optimization method becomes just obvious in such cases where the induced voltages UV can only be
limited sufficiently by earthing the pipe-lines
at more than one point.
[5] CCITT: Directives concerning the protection
of telecommunication lines against harmful
effects from electricity lines.
The International Telecommunication Union
1963
-
POTENTIALS
BURIED
269
AND CURRENTS
CABLE EXPOSED
50
-
12
ALONG AN EARTHED
TO ELECTROMAGNETIC
EFFECTS
OF A POWER LINE UNDER FAULT CONDITION
W.Machczydski
Technical
University
of Poznafi
Poznad,Poland
Disregarding the influence of the
current flowing to soil through the
earth electrodes on the cable subjected
to electromagnetic
effects of a nearby
a - c transmission line may lead, in
certain conditions, to improper results
in calculating the potential distribution.
The paper presents the method of
calculating the potential distribution
along the earthed underground cable,
taking into account the additional
conductive influence of currents flowina
from the earth electrodes of the cable a
on the potential and current distribution.
The method is illustrated with
examples of calculations.
INTRODUCTION
The classical EMC problem in wire
telecommunication
is that dealing with
disturbances caused by electricity
lines. The resulting effects of electromagnetic interference of overhead
high voltage a - c power lines can
range from noise on communication lines
to equipment damage or even personnel
hazards.
The growing use of a - c transmission power lines located in joint use
right-of-way or located in close proximity with buried cables and the increasing levels of voltage and current
capacity make the determination of potentials excited along the cables an
important task.
Where calculations indicate that the
possible hazard could exist, the precautionary measures should be taken to
minimalize the potentials on underground
cables subjected to electromagnetic
effects of a nearby a - c transmission
lines.
As is known, the use of properlydesigned earthing systems permits the
maximum mitigation of cable potentials.
However, in calculations of currents
and potentials along the earthed
underground conductors usually the
additional conductive influence of
currents flowing through the earth
electrodes on the protected earth
return circuit is not taken into ac_
count. Disregarding this fact can lead,
under certain conditions, to improper
results of evaluation of the zone of
dangerous potential on the earthed
conductor.
The purpose of this paper, which iS
continuation of considerations
given
in [31, is to present calculations and
formulas applicable to the analysis of
shield potentials and currents excited
along an underground earthed cable, by
50-Hz a - c power transmission line
sharing a joint right-of-way. The calculations take into account the additional conductive efect of currents flowing
through the earth electrodes on the
potential distribution along the cable.
It is assumed in the paper that the
earth is a homogeneous, isotropic
medium of finite conductivity, that the
underground cable is infinitely long
and that the system considered is
linear.
It is also supposed that the currents and potentials vary with the time
as exp (jut). Therefore the alternating
component of the short-circuit current
is taken into account.
GENERAL
EQUATIONS
In the analysed system, shown in
Fig.1, part of the current flowing
along the buried cable enters the earth
through the connected earth electrodes.
Since the cable is in the current
field of the earth electrode, a part
of the current flowing from the earth
electrode to the earth flows back to
the cable.
Currents and potentials along the
cable are to determined as the superposition of two additional states
[2,3,4], that is:
- current energisation of the cable
by the current leaving the cable
in point x = xk (k = 1,2,...,n),
- conductive energisation of the cable
by the current flowing from the earth
electrode located in point x = xk
(k = 1,2,...,n).
- 270 -
The current 1,k flowing to the
earth through the earth electrode is
determined on the basis of Thevenin's
theorem, hence
VT(x,_ )
lek(xk)
Fig.
1: Infinitely long underground
earthed cable
Hence
I(x) = IO(x)+k;l
v(x)= vO(x)
Ii(x) + k$
+ '; v;(x)
k=l
1; (x) (1)
n
z vi
k=l
+
(x) (2)
where index "0" means the primary
current and potential, that is the
current and potential excited along the
cable when the earth electrodes have
been disconnected,
indices "1" and "2"
- current and potential along the cable
for current and conductive energisation,
respectively.
The shield currents and potentials
excited on the buried cable by an
incident electromagnetic
field can be
calculated using the distributed
source transmission-line
anal sis
technique discussed in [1,2,5 r .
For the current energisation we have
I
I;(x)= - sign(x-xk)+
V:(x)=
ZOIek
2
e
e
,
-YlX-Xkl
and for the conductive
-ylx-Xkl
=
Z + Z K
ek
ck + 'ink
(7)
’
where Zck is the impedance of the cable
joining the earth electrode with the
cable under consideration,
Zink - the
input impedance of the earthed cable
when the earth electrode k has been
disconnected - Thevenin source impedance
and VT(xk) - Thevenin source voltage.
The voltage between cable sheath and
the cable conductor is obtained from
the relation [21
u(x)=
dI(v)
“P
zS
2v
_m
7
_y
e
Ix-v I
C
dv,
C
(8)
where Z, - internal-surface
impedance
of the cable sheath, yc - propagation
coefficient of circuit involving cable
insulation and cable conductor, and the
current I is given by eqn.(l).
EFFECT OF SHORT-CIRCUIT CURRENT
POWER LINE ON THE EARTHED CABLE
IN A
One of the methods used to protect
earth return circuits against the
electromagnetic
effects of a power
line is the earthing of the circuit
under protection in these places where
the highest values of potentials take
place, e.g., in points opposite to the
ends of approach section of a power
line, as seen in Fig.2.
(3)
(4)
\
energisation
(5)
Z
where
v
Z
el
e2
:
- characteristic
zO
cable,
impedance
of the
Y
- shunt admittance
of the cable,
Y
- propagation coefficient of circuit involving the earth and
cable sheath,
0
- earth conductivity,
S
-
distance from the earth
to the cable,
Y!rQ - Sunde's
functions
[5].
electrode
Fig.2:
Underground cable earthed at
points of maximum inductive
influence of a power line
A more general case of the influence of a short-circuit current of a
power line on a nearby underground
cable is shown in Fig.3.
In the system presented in Fig.3 the
fault current IO of a power line influences both inductively and conductively
on a nearby cable.
- 271
5012
-
It should be pointed out that in
case of the system presented in Fig.2,
that is, the case of inductive influence of a power line, the primary currents and potentials along the cable
under the influence are drawn from rewith the ommission
lations (9) - (ll),
of the terms containing functionsyand
R (representing the conductive influence of a power line).
EXAMPLES
Fig.3: Underground earthed cable in
the vicinity of a - c power
line with earth fault
To simplify considerations,
it has
been assumed that the power line consists of only one overhead conductor
with the resultant earth fault current,
while the fault current flows into and
out of the soil through point earth
electrodes placed on the surface of
the earth.
If the line is energised at the
point x = L, and the earth fault occurs
at the point x = 0, the primary current,
potential and voltage between the cable
conductor and the sheath of the cable
take the forms [1,2,51
z
I
IO(x)= +
-YlXi
[sign(x)(l
? .
11
(1 - e
- sign(x-L)
+ 2
wyx,w
-ylx-LI
)I
-Y[v(x-L),
z12z010
vO(x) =
)+
- e
+
usI>
(e -ylx-LI
(9)
-Ylxl
I+
-e
2z11
+ g-
{n(vx,us)
z12Zs10
uO(x)=
2Zll(Y2
_ e-Yclx-Ll)._e
+
YZslo
2nr
-R
-
Y
[ g
-Yi)
(e -yc
Y
rYSl-
{Q(YX,YS)
-+(v,X,Y,S)
n[u,(x-I;), Y,SlH,
(10)
Ix'+
c
-ulxl+
(Y2-Yc2)
[y(x-L
YSI)
R[Y(x-L),
-
e -ylx-LI]+
-t
-t
(11)
- self impedance of cable
212 - mutual impedance
betwee; overhead conductor and sheath
of the cable.
Finally, the resultant current, potential and voltage distributions along
the buried cable being earthed through
earth electrodes, due to electromagnetic effects of a nearby power line,
may be obtained according to eqns (l),
(2) and (8).
;;;=FhZ1l
A 6 cm diameter cable, 0.5 mm thick
tubular copper shield is subjected to
a unit current in a power line. The
cable is earthed at points x = 0 and
x = L through the earth electrodes
located at distances 2 m from the cable'.
The horizontal distance between the
cable and the power line is 10 m.
Height of the overhead conductor of the
power line above the earth's surface
is 10 m.
The remaining data concerning the
parameters of the cable are as follows:
zi = 0.184 + jO.003 R/km, Zs = 0.184
R/km, unit-length capacitance between
sheath and conductor of the cable
- C = 0.1 pF/km, unit-length leakage
conductance of cable insulation between
sheath and conductor of the cable G = 0 S/km.
Results of calculations of potentials and voltages due, in the buried
earthed cable, to electromagnetic
effects of a power line are shown in
Figures and Table (moduli per unit
current).
Calculations have been carried out
for the case of earth fault occuring
in the close proximity of the cable
(conductive and inductive influence of
power line) - Fig.4, and for the case
of inductive interference only (the
earth fault is remote from the approach
section of the power line) - Fig.5.
In Fig.4 the potentials are plotted
as function of x, whereas in Fig.5, the
influe,nce of cable shunt conductance
on the maximum value of potential of
the cable is shown.
In Figs 4 and 5 the curves IV'1
represent potential for the case where
the additional conductive effect of the
earthing currents was not taken into
account.
The influence of earth conductivity
and shunt conductance of the cable insulation on the values of maximum voltages (at the point x = 0) between sheath
and cable conductor is shown in Table.
In the Table entry with prime corresponds to the case where the additional conductive effect was disregarded.
CONCLUSIONS
The results of calculations show
that the earthing of buried cable reduces significantly the value of potential excited by electromagnetic
effects
of an a - c power line. The values of
Potentials obtained on the assumption
- 272 -
IVI
hV1
13(-j
-0,s
0
Fis.4:
0,5
Potential
I,0
I,5
distribution
2,O
along earthed
2,s
underground
3,0 h-d
cable
that no additional conductive influence
exists may be smaller by 20% (for
maximum values).
The influence of the additional conductive effect on the potential distribution along the protected cable becomes obvious in case when the cable
having an insulation of great conductance is buried in the earth of low
conductivity.
The earthing of the cable may result
in increasing of voltage between sheath
and cable conductor, which is noticeable for well insulated cables.
The influence of the additional conductive effect on the voltage between
the cable conductor and the sheath
practicaly does not exist.
IVI
hV1
300
0.1
1.0
REFERENCES
Gi [ S/km1
10
[ll.
Fig.5: Maximum potential vs cable
shunt conductance for various
values of earth conductivity
Table. Maximum voltage between
and cable conductor
cr=5.10-3S/m
sheath
cr= 10B2S/m
( IUl / mV
1
90
86
83 1 90
85
79
1
1 IU'II mV
/
95
91
841 93
87
79
1
Krakowski,M.: Currents and potentials along extensive underground
conductor.Proc.IEE,Vol.llS,No
9,
1299 - 1304 (1968).
lI21.Krakowski,M.: Obwody ziemnopowrotne. WNT,Warszawa,
1979.
Electromagnetic
[31. Machczydski,W.:
effects of a - c transmission lines on extensive conductor earthed through impedances. Seventh International Wrockaw Symposium on
EMC,WrocZaw, June 18-20, 1984
485-494.
J.W.: Tieorija i ras[41. Striiewskij,
czet wlijanija elektrificirowannoj ieleznoj dorogi na podziemnyje
metalliczeskije
sooruienija.
Izdat.Lit.po Stroitielstwu,
Moskwa, 1968.
[51. Sunde,D.E.: Earth conduction
effect in transmission systems.
Dover Publication, N.York,1968.
-
273
5113
-
COUPLING AND PROPAGATION
OF TRANSIHNT CURRHNTS ON MULTICONDUmR
TRANSMISSIONLINE?3
J.L. ter Haseborg*, H. 'Winks*, and R. Sturm**
*TechnischeUniversitatHamburg-Harhurg
Hamburg Germany
**NRC Defense Research and Development Institute
Munster Germany
In
order
to
estimate
the
protection
efficiency or to realize an opt.imumprotection
for sensitive electronic devi~cesrespectively
by special protection circuits aqainst guided
transient currents - e.g. caused by liqhtning
(~23) or NEMP - the variation of the time dependent or frequency-dependentshape of the
interfering currents by the transmission line
has to be known. Particularlythe edge steepness of the surqeinfluencesthe response of
ft;?yl protection devices, e-q. gas arresters
tronsmissio n -
protec4 on
circuit
.----
!
,
L----
I -----J--f*--’
electronic deviceNine tefmination) to be protected A
i-._._._
A shielded multiconductortransmission
line is considered.The coupling process
between cable sheath currents and conductor
currents as well as the propagation of sheath
and conductor currents are described analytically. Starting from this description a
computer code is developed.
Introduction
There are two applicationsfor the cornputations concerning coupling and propagation
of transient currents on multiconductortransmission lines. The first application is explained by Fig. 1.
The coupling and propagationof incoming
pulses,runningto the input terminals of protection circuits as well as the propagation of
residual pulses at the output terminals of responding circuits, running to the input terminals of the electronic device to be protected, are of special interest. Referring to
Fig. 1 the shape, particularlythe edge steepness, of the pulse@determines the response
(dynamic threshold voltage) of the protection
circuit. Often the electronic device to be
protected and the protection circuit are not
directly interconnectedbut separated by a
line of the length 1 as shown in Fig. 1.
Starting from a definite residual pulseOat
the output terminals of the responding protection circuit the transmissionline constants in connectionwith the line termination
dertermines the total pulse@at the input terminals of the device to be protected.Assuming
the worst case the amplitude of the total pulse
@nay be two times larger than the amplitude of
the pulse@ .
The second application,which is also
typical for many cases in pracitce, is shown
in Fig. 2.
F’ig.
1:
I
*
._._or LRMP-induced
and residual pulses@on
transmission lines
Propagation
pulses@
of
NEW-
EMP
termination
1
input terminals 1
S
termination
.-.-.
2
input terminals 2
Fiq. 2: Transmission line terminated at both
ends with arbitrary impedances
A multiconductortransmissionline (length 1)
is terminated at both ends with arbitrary impedances. When the line is excited by an
electromagneticfield (NEMP or LFMP) the total
currents I
Cl1 *.* ICln Or *c21 *.* ICZn
respectivelyat the input terminals of the
termination 1 or 2 respectivelyare of interest. However,quiteoften these terminals are
not accessible and therefore it is impossible
to measure the voltages or currents respectively at the pins of the terminations.Starting from an induced current as lumped source
on the cable sheath the computer code developed allows the calculationof the total
currents at the input terminals 1 and 2 assuming arbitrary terminations.
In this paper results concerning the
second application of this computationprocedure will be presented and discussed. The
first application,thatis coupling and propagation of induced pulses on transmission lines
as well as propagationof residual pulses,
caused by responding protection circuits (s.
-
274
-
d1
-_=-y
'* u
dz
-
Fig. l), will be the subject of another publication.
Theory
basis for the calculationsis the
transmission line theory, particularlythe
transmission line equations in the frequencydomain. This theory is applied on the complete
transmission line, that means on the inner
conductors as well as on the cable sheath
(shielding).Generally it is not possible to
describe the sheath currents e.g. the currents
on a braided shield by means of the transmission line equations. Starting from the
transfer impedance which describes the coupling between sheath currents and inner conductor currents an equivalent cable sheath has
to be found showing the same frequency-depndent transfer impedance as valid for the real
sheath and which allows the application of the
transmission line equations.
A
(4)
the transition to ntl conductors provides the
equations for multiconductortransmission
lines in matrix notation:
(5)
d [II
- dz = -[y’[
Q’]
.
(6)
The insertion of (6) in the derivative of (5)
yields the wave equation for multiconductor
transmission lines:
d2[El
= [z’ 1 [y’ 1 [El
l
Fig. 3 contains e.g. for a braided shield
typical curves concernin the tran f r imp?3_)have
dance [2]. Computations7 s. Kaden BeI
shown that it is possible to realize a cage as
in Fig. 4, showing a freguencyrepresented
depndent transfer impedance,which is largely
identicalwith the transfer impedance of the
real sheath.
l
(7)
dz2
Concerning a definite braided shield, Fig. 5
shows a good agreement between the transfer
impedance of the real sheath (solid curve
number 1) and the transfer impedance of the
equivalent cage (dashed curve).
The values are dependent on
- diameter of the cage
- number of cage conductors
- diameter of the cage conductors.
This transformation:
real cable sheath + equivalent cage
enables the applicationof the transmission
line equations not only on the inner conductors but also on the cable sheath. Referring
to the equivalent circuit, shown in Fig. 6,it
is no problem by means of the transmission
line theory for multiconductortransmission
lines, besides the considerationof inductive
and capacitive couplings, additionallyto take
into account the line losses, that means ohmic
losses in the conductors and in the sheath as
well as frequency-dependentlosses in the insulation.
Generally the transmission line constants
R', L', M', G'and C' - referring to Fig. 6 which are the elements of the matrix [z'] or
[Y'] respectively,are measured. In thys case
&ese constants are computed according to [4],
[5].
Starting frcnnthe transmission line equations
for a two-wire line in the time-domain:
au
xi= -
a
i
$
u
(R'+ L' -&
ai.
,,a
YiZ= - (GtC
(2)
or in the frequency-dcanain
respectively:
dV
z=-z
'. I
(3)
2
2
IO0
10“ 2
lo0 2
5
10' 2
MHz
10'
fFig. 3: Typical curves showing the transfer
wance
of braided shields [2].
co: optical coveraqe
5
By means of this multiconductortransmission
line theory an arbitrary nuker n of lines
can be considered. In (7) the voltage of one
line is coupled with the voltages of all other
lines. In order to solve this coupled differential equation system a linear transformation
provides decoupled wave equations for multiconductor transmission lines:
[El = [VI- [WI ,
(8)
5113
275 -
The matrix [v] has to be chosen in such a
manner that the insertion of (8) in (7) results in decoupled differentialequations.
Insertion of (8) in (7) and multiplication
with [VI-'
provides:
-
[g’ ] [y’ I [VI
= [VI Cr21
This equation represents an eigenvalue problem [6]. The decoupled differentialequation
system is shown in (12):
d2[W]
__
= [r'l [WI .
dz2
In order to obtain for [W] decoupled differential equations it is required:
[g-l [g’J[y’I[vl
= [r21
(lo)
whereby [r'] represents a diagonal matrix.
'(11)
(12)
The components of [w] are designated as natural
waves of the multiconductortransmission line.
'The square roots of the elements of matrix
[r'] are the propagation constants of the
natural waves:
(13)
Yu = au + jfiu
whereby a or (3 respectivelyrepresents the
attenuatign con&ant or phase constant res_u. The general
pectively of the natural wave W
solution of the wave equation contains an incident and a reflected guided wave:
able
‘al 1
inner
shield)
consist
conductors
Fiq. 4: Cage conductors as equivalent cable
sheath concerning the transfer impedance
The uantity z marks the location on the line,
and i!
W.(O)] or [W (O)]respectivelyare the incidenEior reflec& waves respectivelyat z=O.
By means of equation (14) and various
matrix operations the equations for multiconductor transmission lines can be obtained:
for the line currents the following expression
is valid:
This formula shows clearly the two parts belonging to the incidentorreflected wave respectively. Generally it is not easy to cqute
the matched termination for a multiconductor
transmission line, because terminating impedances between all conductors are necessary.
In case of the computationprocedure presented
by means of various matrix operations the
matched termination can be found. Matched
terminationmeans, no reflected waves are
existent, therefore the following equation is
valid:
[v]-‘*[u(O)]-[r]-‘*[rs]-‘*[~’
]*[&(O) ]= 0 .
(16)
After various mathematicaloperations the Ymatrix for the matched terminationof a multiconductor transmissionline is obtained:
I&J=
fCnzl
Fig. 5: Transfer impedance (normalized),
- solid curve: braided shield
- dashed curve: equivalent cage
Equation (lo), multiplied with [v], provides
(11):
C3’ l-l~[~l~[rl~[~l-l
,
(17)
Computation results
In this paragraph results - referring to
Fig. 2 - are presented.As already mentioned
above problems and results - referring to Fig.
1 - will be the subject of another publication.
Concerning Fig. 2 a multiconductortransmission line is excited by a spatially short,
pulsed electramagneticfield. In the present
case the computationsdo not start from this
field but from a sheath current Is, which can
arbitrarilybe assumed, as lumped source lo-
- 276 -
a)
termination
1
termination
2
600R
p@]g
600R
b)
piq. 6: FQuivalent circuit of multiconductor
transmissionline of infinitesimal
length dz
cated at the line center. The parameters of
the dissipative transmission line investigated
show the following values:
- lengthl=lcom
- 4 inner conductors
- transfer impedance of the sheath,servingas
shielding, according to Fig. 5 curve no. 1
- the puls-shaped sheath current I assumed
has a rise time t =50 ns and a h&f-amplitude pulse duratign tf=5,5 us.
These values are valid for figures 7 and 8.
Referring to Fig. 2 the curves of Fig. 7
show the currents I_
(termination1) and
termination
ICI
1
ll!flI
l/L_
1.1
85
0
-.OS
0
2
i
6
tcps1
0
2
4
6
termination
t bsl
2
G
.05
0
L------
2.1
-.05
0
2
L
6
tIpsI
0
2
L
6
tIpI
0
2
1
6
t&4
a)
terminations
1 and 2
600R
600R
b)
termination
4
termination
0
Fig. 8
1
4
Fig. 7/%:
10
c
20
30
t lps1
-.L
D
10
20
30
tIpsI
Coupling and propaqationof a pulse
(t,=50 ns, tp=5,5 ps) on a line ter-
minated at both ends according to
Fig. 2, a) line terminations
b) 1.1 incident guided wave,
1.2 and 1.3 reflected by
termination 2 at termination 1
2.1 incident guided wave,
2.2 and 2.3 reflected by
termination 1 at termination 2
c) total currents at terminations 1 and 2
(termination2) on one of the four inner
2
IC2
conductors.Both ends are terminated with 6coR
symmetricalas shown in Fig. 7a. In Fig. 7b
the incident and in each case two reflected
guided.waves are representedthat means at
both ends two reflectionsare considered.Fig.
7 c shows the total currents at both terminations. Fig. 8 shows the corresponding
currents for the case that the end"l"is terminated with 6cxQ symmetricaland the end/'2"
is short-circuitedaccording to Fig. 8a. In
figures 7 and 8 the currents are normalized on
the amplitude of the primary sheath current
Fiq. 7:
IS
-
Apart from the transmission line COnstants the pulse shapes are dependent essentially on:
- line length
- line terminations.
The pulses are largely characterizedby line
resonances which are dependent on the line
length. In [7] a similar example has been ccmputed, that is a lossless transmission line
consisting of two conductors excited by a half
sine wave of width 20 ns.
Conclusions
Here only a small number of results can be
presented. We have carried out a lot of canputations and besides line length and line
terminationswe also have varied the primary
sheath current pulse I particularly its rise
time. The smaller the &se time the larger the
pulse coupled into the inner conductors.Responsible for this effect is the frequencydependent transfer impedance.Referring to the
transfer impedance shown in Fig. 5, taken as a
basis for the computations,particularlyrise
times less than approx.severalhundred nanoseconds cause comparativelyhigh pulses on
the inner conductors.
The research work concerning coupling and
propagation of transients on multiconductor
transmission lines is going on.
References
[l] ter Haseborg,J.L.;Trinks, H.: Protection
circuitsforsuppressing surge voltages
with edge steepness up to lo KV/ns.
277
5113
-
5th Symposium on electromagn.Cornpat. e,
Zurich, March 8-10, 1983
[21 Homan, E.:
Geschirmte Kabel mit optimalen
Geflechtschinnen. NTZ, Heft 3, 1968
und Schirmung in
[31 Kaden, H.: WirbelstriSmE!
der Nachrichtentechnik.Springer-Verlag,
Berlin, Gijttingen,
Ileidelberg,
1959
[41 Clements,J.C.;Pau1,C.R.; Adams,A.T.:
Computationsof tlX?capacitancematrix for
systems of dielectric-coatedcylindrical
conductors. IEEE Trans. Electromagn.
ccXnpat.,vol. FMC-17, no. 4, Nov., 1975
of
[51 Pau1,C.R.; Feather,A.E.: Ccanputations
the transmission line inductance and capacitance matrices from the generalized
capacitancematrix. IEEE Trans. Electromagn. Ccanpat.,vol. W-18, no. 4, Nov.
1976
[61ter Haseborg,J.L.;Trinks,H.:
Transient
response and protection of multiconductor
transmission lines. InternationalAerospace and Ground Conference on Lightning
and Static Electricity,Orlando, USA,
June 26-28, 1984
[71 Agrawa1,A.K.;Price,H.J.;Gurvaxani,S.H.:
Transient response of multiconductor
transmission lines excited by a nonuniform
electromagneticfield. IEEE Trans.
Electromagn.Ccmpat., vol. EMC-22, no. 2,
May 1980
-
279
52
-
14
Response of a S it-w
le-ConductorOverhead
Wire Illuminated by an InhomogeneousPlane Wave
-
-
F. PALADIAN, J.P. PLU?4EY, D. ROUBERTOU, .I.FONTAINE
University of Glermont-Ferrand
France
1. Introduction
In this paper, we obtain the time domain
response of single conductor overhead wire illuminated by an inhomogeneous plane-wave. For
this purpose, we develop an E- integral equation formulation for the current in the frequency domain and discuss a numerical procedure used to solve this integral equation,
based on the application of the method of moments and the finite difference technique.
This method presents several advantages
over he transmission line theory : first,
coupling between horizontal and vertical wires
is not taken into acount with this theory, and
then, results from transmission line theory
fail to exhibit the resonances for the structure.
At last, a Fast Fourier transform (FFT)
algorithm is used to convert the frequency
domain results into the time domain response
Numerical results are presented for the current induced on the structure placed over
perfect g.round.
2. General formulation
Figure 1 shows an overhead single conductor, horizontal with two vertical terminations penetrating the ground. The angle between the vector 2 for the plane-wave and the
horizontal line is $.
Regions 1 and 2, respectively the soil
and the air, are characterized by (cl = eIcO,
1-11
= 1-I,,
oI) and (e2 = Ed, u:! = u,, u2 = 0)
where E and po are free-space parameters.
0
we consider "thin wire hypothesis" that is to
say antenna radius is much smaller than vertical and horizontal parts lengths.
We consider an observation point P, with
c2ordinates (X, Y, Z), which is labelled by
OP = r.
If we design the Hertz vector potential
by t, the scattered field at P is given by :
1.
- if P region 1 :
it,(;)
= (k:+VV.) I
$1)
6,
+.(
r )d;'
(1)
structure
- if P
g,(G)
region 2 :
= (k$+VV.) I
$2)
+ +.I
(r, r )dg'
(2)
structure
where k represents the wave number :
k? = w2p
- jwpiui
1
i Ei
i = I,2
Designating &he unit normal as fi and the
incident field as El, we obtain the integral
equation :
fi.
(@c:,
+ it(Z))
=
0
(3)
From each equation (1) and (2), we
obtain as many equations as there are vertical and horizontal dipoles respectively in
regions 1 and 2. (for the studies case, there
are two equations if P belongs to region 1
and three equations if P belongs to region 2,
so we obtain five equations).
In equation (l), we consider z component
of $1
in region 1 and the kernel depends on
we consider the source as :
- either a VED (vertical electric dipold
or a HED (horizontal electric dipole) in
region 2.
- or a VED in region 1.
In equation (2), the kernel depends on
we Sonsider either x-component or z-component
in region 2 and it is different when
of E2
the source is
Figure 1 ,: Geometry of the structure
The time variation for all the expressions given in this paper is represented by
the time factor exp (jut).
We take into account following hypothesis : first, vertical and horizontal lines
conductivity is supposed infinite, and second
- either a VED or a HED in region 2.
- or a VED in region 1.
The kernels of equations (1) and (2) are
composed of Green's functions and Sommerfeld
integrals (cf. annexe).
The numerical solution of equations (1)
and (2) is obtained via the application of
-
280
-
the method of moments (2); we used Point
Matching method that means currents are considered as constants along every patch of length
A on horizontal or vertical wires.
Application of the method of moments
allows one to write integral equation in terns
of the following matrix equation :
k: f
Zj+A/2
zj-A/2
+(z
hg>i”j + (zhg).I.
hg
or:Z..=(Z
1J
4
source
term
or
(Zmn)(In) = - (Eim)
(I”)
represents unknown current
matrices.
&
In each sub-matrix, every component is
broken up in three terms : a source term
(when we consider region 2 as infinite), an
image term (when region 1 is considered as
a perfect ground) and a corrective term
composed of Sommerfeld integrals (when region
1 is considered as a lossy ground).
For example, the expression of the incident
field is given when the observation point
is along the horizontal part of the structure
in region 2. Ba;\;osEormalism(2) is used with
the time factor exn
_ (iwt).
_
E;(X,h)+$?o
+ $Q
!; I,W’k&
~hI,(0,z9
+ $$$
x=x
V12dZ' X1=0
Z=h
(G22-G21+k&)dZ'
&
II'Ix(X',h)
($
x=x
X'=O
Z=h
+ k;)(G,,-G,,)
0
x=x
+
a2
k2
2 (w
+ +f$$
+ *22 I
V22
I,’
L&Z’)~
dX’
a2
Z'=h
z=h
/-hrz(L,z,)
&
x=x
“,‘r;
(G2,-G21+k;v,,)dz’
V12 dZ'
where expressions of Green's functions (G22
and G21) and Sonunerfeld integrals (Vl2, V22
and V22) are given in Annexe.
In this expression, every integral gives
us expression of a sub-matrix of the third
line of (4) ; we obtain respectively Zhg ,
Zhg, zhh, Zhd and zhd .
For example, we obtain :
ij
ho
W
Zj+A/2
f
zj-A/2
a2
hg)C
..
! lJ
corrective
term
&
f(X,Z)
=
f(X+A,Z) + f(X-A,Z) - Zf(X,z)
A2
f(X,Z): =
f(X+A/2,z+A/2)-f(X+A/2,2-n/2)
A2
_ f(X-A/2,Z+A/2>+f(X-A/2,2-n/2)
A2
The first purpose of this paper is to
obtain the current value at the point M in
the frequency domain.
Then, these results are used to investigate the transient behaviour of the antenna
current mainted over perfect ground.
As a first step, the transfer function
(impulse response) of the structure current
is computed using the previous results. This
transfer function is then multiplied by the
spectrum of the EMP (electromagnetic pulse),
and finally the Fourier inversion is performed numerically via a Fast Fourier Transform
(FFT) routine to obtain the transient response
E(t)
= El(e-et- eeBt)
and by a Fourier transform, we obtain :
E(f) = E,(&
- &)
- j E1(;&-&'&
;I;"
0
zhg =
Xi
EMP form is given by :
x=x
+ *
) lJ
image
term
=
;': ;
By means, it is possible to reduce
calculation times by taking into account all
symmetries of each term.
To avoid difficulties that come from
differential operators inside the integrals,
we have chosen the finite-difference scheme
for performing the differentiation outside
the integrals. The two following formulas
are used :
(zmn) represents the impedances
matrices.
where
X
a2
V22 dZ'
axaz
x=x;
axaz Gz2dZ' X'= 0
Z=h
At a point M on the structure (cf.
figure 1) the current transfer function may
be defined by H(f), which is the deltaresponse of the structure at point M. The
transfer function can be constructed discretely from previous results. Using the superposition theorem, the transform of the current
at point M due to the EMP E(t) can be written
as :
I(f) = E(f).H(f)
and the transient response of the structure
current at point M can be expressed as :
4-m
i(t) = 1
I(f) ej2xft df
-
281
3. Results
a) In the frequency domain :
Results are given for a case closely
related to Taylor and Castillo's (4) with a
different height of horizontal part to take
the same patches length on horizontal and
vertical part of the structure.
We used 40 patches on horizontal part
and 1 patch on each vertical part : this choice
ensure that moment method solution converges.
Response in the frequency domain of the
structure is given for a perfectly conducting
ground, when vertical parts are each loaded
first by 100 R, second by the caracteristic
impedance Zc.
52
-
14
lines theory is equal to zero that is verified between resonance frequencies (5) *
III
(Al
,
4.p
2f.QMh.z
3.19
2.18
ld?
zc = 120 Log ($) = 635 n
First, currents are computed along the
structure for a frequency corresponding to a
wave length equal to the total structure length
(figure 2). The plane-wave incidence is 90'.
So the excitation of the structure is symmetric
about X = $ ; theLcurrents are of course symmetric about X = 2.
Figure 3 : Response in the frequency domain at.
point M
Figure 4 : next page
b) Time domain results.
Previous results are used to obtain the
time domain response of the structure at
point M.
The structure transfer function H(f) is
computed at the total of 1024 frequencies
in the range of :
0
Figure 2 : Response in the frequency domain
along the structure.
Limits of moment method are given by
L > X/6
These conditions are directly linked to
the basis functions choice and they ensure
results convergence ; they have been specified
by many testings.
For this case, we may obtain results
for :
3 MhZ < f i 40 MhZ
Currents are computed at point M for
this frequency range.
Resonance frequencies are given by :
where
<
f < 40 MhZ
With this choice, all the sharps peaks
are taken into account.
f = 3 MhZ represents the lower limit
of moment method so transmission lines theory
is used for 0~ f < 3 MhZ and moment method
for 3 MhZ < f < 40 MhZ.
The transform of the EMP is then multiplied by the structure transfer function to
obtain I(f). Note that at f = 40 MhZ, the
resultant frequency domain currents have
decayed sufficiently so that zeros can be
added for f > 40 MhZ.
Now that the entire frequency domain
is constructed numerically, a Fast Fourier
Transform routine is employed to obtain the
time domain response i(t).
The ringing effect linked to the sharps
peaks of the frequency spectum makes plotting
i versus time response difficult. So, figures
5 and 6 respectively shows the time domain
response only up to 1,25 us and the envelope
along the all range of times.
Ii1 (A)
800f
700..
800.
500'
fr = mcl2L
400.
c is the velocity of light
m=l ) 2, 3, . ..
300.
200,.
In this case, for I$ = 90", the resonant
current modes corresponding to odd values of m
are not excited : the excitation of the structure is symmetric about X = L/2 and these resonant current modes are not excited (figures 3
and 4).
Current value given by the transmission
100
0.75
’
1
Figure 5 : Time domain response at point M
for 25 ns ,< t,< 1,25 us
-
282
-
W
-y2(z+z')(Y2-y~)
=2fme
22
W 21
k:Y, + k;Y,
J
(A,->
Ad),
'
(Y2-yl)Jo(Xr) AdA
=
-Y2(z+z')
U
200
'-I
Jo(Xr) Xdh
22 =
Yl+Y2
0
100
BW22
Figure 6 : Envelope of time-domain response
at point M
4. Conclusion
Our purpose is to obtain time domain
response of the structure by using Fast
Fourier Transform.Frequency domain results
show that it is necessary to resort to an
integral formulation of the problem. Transmission lines theory actually fails to exhibit
the resonances for the structure that would
affect time-domain results computed by FFT.
Results obtained for a perfectly conducting ground : this case is unrealistic but
represent a step in the solution to the problem when the ground is imperfectly conducting.
a) Green's functions
.-jkzR2
.-jk2Rl
G22 = ~
G21 =R2
e-j
w21 =
we have
+
G12 =-
e-j
x-2 = x2 f
y2
order
References
11) A.J.
tion
Poggio, E.K. Miller, 'Integral equasolutions of three-dimensional scattering problems', in Mittra, R.(Ed.) :
'Computer techniques for electromagnetics' (Pergamon, 1973), pp. 159-263.
(21 R.F.
and h = 1~')
(4)
m
Jo(Ar) XdX
0
k;Yl + k:Y,
co
"21
eY1z-Y2z'
321
Jo(hr) XdX
0
kZY, + k;Y2
e-Y2z+Ylz'
"12 = 2 jm
0
V
11
L 2
Jrn
0
i = I,2
Jo represents Bessel function of zero
(3)
b) Sommerfeld integrals
"22 =2j
ki)l’:!
Harrington, 'Fiel Computation by
moment methods', Ed. Mac Millan,
New York, 1968.
RI = (r2+ (z + h)2)1'2
where :
(X2-
*
with : R2 = (r2+(z - h)2)1'2
and
:
r represents horizontal distance from
the observation point to the origin 0.
kl RI
RI
=
"21
*
Gll =R2
Yi
(& - &'i
with real (yi) > 0
Rl
kl R2
= k; V22 - U,,
k2,W2, = -$ (2G21-(k;+k;) "22)
Annex
Expressions of Green's functions and
Sommerfeld integrals used in Ba?los's formulation.
az
(5)
A. Bazos, 'Dipole radiation in the presence of conducting half-space', NewYork : Pergamon Press, 1966.
C.D. Taylor, J.P. Castillo, 'On e'lectromagnetic-field excitation of unshielder
multiconductor cables', PEEE Transactfons
on electromagnetic compatibility, vol.
EMC-20, NO. 4, November 1978.
A. Albert, Jr. Smith, 'Coupling of external electro-magnetic fields to transmission lines', John Wiley and Sons,
New-York (Eondon) Sydney (Toronto,1977).
Jo(Xr) Adh
k:Y, + k;Y1
eYl(z+z')
Jo(hr) XdX
k; Y2+ k;Y,
Figure 4 : Response in the frequent domain
af point M.
-
TIME DOMAIN
SCATTERING
283
5315
-
BY THIN WIRE STRUCTURES
A HOMOGENEOUS
F. MAUMY, B. GECKO et 0. DAFIF
Universite
de Limoges - U.E.R. des Sciences
de Communications
Optiques
et Microondes
(LA 356 du C.N.R.S.)
123, rue Albert Thomas - 87060 LIMOGES CEDEX
Laboratoire
ABSTRACT
This
paper
electromagnetic
shows
pulses
of any
form,
terized
by a conductivity
the
above
capital
current
stage
induced
equations,
the
by
thin
o.The
is
the
on the wire,
taking
scattering
wire
a homogeneous
into
of
problem
from
is resolved
by integral
in
-Figure
which
of
the
k’
charac-
determination
account
The
space-time
presence
space
of
Some
I-
applications
domain
of
and electric
are
chosen
in
telecommunications
or railway
systems.
theoretical
to
responses
through
on these
installations.
perfectly
was
treated
wire
illumination
of induced
currents
scattering
problem
structures
[ll,
of a perfectly
[2],
with
takes
a finite
into
account
conductivity.
Fresnel’s
approximation,
in
integral
equations
verified
induced
on
wire
[41
and
structure.
the
the
simulator
as antennas
earth,
with
The
of
method
of
interpolation
is also applied
radiation
or essay
transmission
of
waves.
moments
used
in pulse
emission
11.1. Principle
the
P
the
any
in the
(I?“,
Gal,
the
superposition
at a point
(gd,
of
a wire
perfec-
half-space,
t which
is respon-
Ad). The field
is
the
M of the
it
(?$I)
expressed
on
:
by
(2)
(3)
determined
electric
half-space
conditions
called
as
the
half-space
Knowing
@,
field
a current
operator
wire
arrives
wave
half-space,
in
= L[I(M,t)l
ting
I%
(1)
Ed(P,t)
boundary
in
it’s written
integral
the
obstacle,
+ Ed(P,t)
equations.
point
(I?,
half-
this
to
field
Maxwell’s
larly
total
structure
= Ea(P,t)
an
wire
@“, fia) illuminates
of
with
the
a reflected
point
due
on its surface
field
by
8 radiated
and therefore,
wire
surface,
on the
the
at
particuone applies
perfectly
conduc-
:
in presence
pulses
s’;‘?(M,t) = 0
where
s
PROBLEM
of the method
any
in
+ Er(P,t)
wave
(4)
is the
at the point
II - THEORETICAL
wave
At
of diffracted
is
incident
is
= Z’(PJ)
point
L
Without
creates
,
treated
earth.
it
conducting
sible
is
Therefore,
current
are
to some
electromagnetic
line.
a
establishing
solution.
This work
problems
by the
Lagrange
to the numerical
field”,
a
It consists,
the
the
obstacle,
QP,t)
paper
1).
“applied
induces
conducting
plane
without
tly
[3J,
the
(figure
half-space
problem
ground,
When the
ground.
ground
fir)
Ea(P,t)
structures
pulses
Transient
in presence
This
in
of
the knowledge
conducting
or dielectric
real
work
electromagnetic
proceeds
by
the
the
both
INTRODUCTION
The
of
the
equip-
above
when
on
protection
( &?I
Iair
P,,rr
,r,,,r,,,,,,,,,,,,,,,,
earth
the
the earth.
ments
:;‘:y_e&+
structures
ground
established
directly
domain,
ABOVE
GROUND
Using
lows
to
unit
tangent
vector
to
the
path
M.
obtain
(2) and (3), this
an
integral
condition
verified
(4) alby
I(M,t):
-
z.{L[I! M,t)] + ca(M,t))
11.2. Applied
field,
First,
in
we
the
the
order
time
the
reflected
our course
of action
is the
following.
consider
the
reflection
domain.
domain
by aninverse
the
air/soil
we
use
Laplace
the
R(t)
takes
and
the
function
the
earth
we present
the
= $$
I
(e-d)
is
at
e-T,
(ii)
- 02
_(6-d)
the
horizontal
-t
d
,
+ (8-d) IO $1
with
e
where
tion
-- Bt
e
*
0‘;
B=
CI
E E r
sin’
obtained
Thus,
consists
of
two
the dielectric,
of order
RTM(t)
reflection
parts,
the other
0
2 shows
dence
angle
of
the
half
one
of
which
R(t)
is
due
to
(8)
the dependence
is
= E’(o,t)
11.3. Integral
The field
poles
on the
the
computed
at
any
from
point
the
P
plane
waves
gd = L(I) being
boundary
conducting
the
wire
integral
condistructure
equation
verified
1:
to *(t) et T*(t) are vectors
on
the
earth,
[T*(t)]
= [R(t)l.[6*1
where
(To, d o*) are respectively
the source
dipole
R(t)
is the
by
the
which
express
connected
with
[R(t)] [5], [6] by :
matrix
= [R(t)l.[:o*l
the unit
and to the image
reflection
dipole,
matrix
of the
connected
vectors
to
dipole.
field,
with
radiated
the
functions
sin 4’
o
II61by :
et RTM(t)
[R(t)].
- RTM(t)
cos
- RTM(t)
sin 4’
i
the
(M~,A,P)
by adding
of all the electric
. Frequency
have
by
domain
using
the
\iq*( image)
- RTM(t
angle
of
incident
of the Fresnel’s
been
validity
obtained
comparison
ground,
condition
:
zones
Fresnel’s
plane
coefficients
[7]
with
of
reflection
in
the
approached
coefficients
frequency
exact
domain
method
using
integrals.
For an
1
o
domain
Sommerfeld’s
‘\
cos $’
3).
The
earth
RTE(t)
0
(figure
wire.
3 -
RTE(t)
method
up
di-
-
polarization
11.4. Validity
is obtained
@’
0
is
method
- Figure
to obtain
reflection
field
6 (t - +G
ed(P,t)
A is suppo-
of R&t) on the inci-
field
dEd(P,Mo,t)
the
3).
of the
[@I
equation
the contributions
perflectly
reflections
4’
(3
one
RTE(t)
at the interface
j$‘(p,t)
on
from
of the
field
application
by the current
conductivity.
reflected
space
be
to the conductor.
and the earth
The
can
function
R(t) = Rc 6(t) + R&t)
Figure
allows
the
and where
polarization.
the
[_5], the
on
the
a tb
Bessel functions
expression
for a vertical
reflected
spring
at the point
coefficient
a
0)
b’f3
e =-
similar
one
at M; (figure
the
at
formed
dT
and 1.
A
by
seems
The diffracted
reflec-
(11 (9)
0(
IO and Ii are the
and
emited
MO, is
. s(t)
a = cos 8;bnJcr-sin*
i
with
is
domain,
e-av2
e-d(t-‘d
_- Bt
2 ] + sb
b*B
d=jyr-g
done
known
polarization.
$
situated
time
at
half-
frequency
(RTE ou RTM).
(@.)
02
1
dipole
which
which
the
used
in
situated
wave
wave
expressed
only
emdt - a?_$-
wave
in
very
consists,
dipole
straight
oneself
the
The reflection
It is written
RTE(t)
image
characteristics
R(t)
a
ground
product
incidence.
for
one
the
(6)
[61. Here,
analytically
tion
of
by
the
P by
seats
earth,
[2]
supposing
sed
into’ account
angle
in
reflected
= R(t) (;) @(o,t)
g’(o,t)
it
transform.
interface,
by the convolution
is given
in
one
the
method
point
coefficient
Then,
above
domain
determine
Near
field
When
space
to
frequency
-
(5)
field computation
In
electric
= 0
284
the
electric
height
dipole,
h must
situated
verify
the
above
following
I
-
0
01.
,‘.‘*‘..‘*“.“’ 150
100
“,
1,
50
,‘,
208
285
5315
-
Er.“,l”..l.““““““’
0
50
250
100
150
200
240
t(ns)
t,(ns)
_d--.5
-5 -
-4
Conduct
I
i vi ty:
lG3
mho/m
t
t
-I
t
t
t
(, . . . .
,.
‘.
.’
-1.5”
_-3
Conductivity:
-2.5 t
5
(*la
mho/m
5
-25 i
)
18
(Xl0
(-
)
50
100
150
(
250
2h0
t(ns)
-rib
-108
-100E
Conductivity:
Conductivity:
18
-2
mho/m
-150
-1500
-200
-250
IL
(
- Figure
ANGLE
Permittivity
OF
2 -
INCIDENCE
E, = 10.
. . . . 60"
--_
---
450
300
00
REFLEXION
ANGLE
AND
OF
FUNCTION
CONDUCTIVITY
INCIDENCE
RTE
(t)
oINFLUENCE
(degrees)
:
18
-1
mho/m
-
-
h>--X
07
Ee = Er +-z-JWEo
JF7;1
from ‘thlis height
condition,
[S] a frequency
0,7)‘cZ
h2
f >
(12)
it is possible
:
condition
using
1
Fresnel’s
available
so
magnetic
pulse
are
reflection
that
all
the
wire
behaves
like
a
is
The
seems
However
first
penalizing
high
For
I
Moreover,
(sod
our
attention
border
effects,
equation
TM
by the
sampled
numerically
method
resolved
sequence
[ 21.
proceeds
in the sample
t-R*/c
The
original
Ru
=
by
of
stage
the
(so,+-
of
ximed
I(so
NS
where
of this
time
$$so,T)
of time
1,
uq
=
I
seen
to
=-2
can
term:
dT (15)
be
intervals
s;+
1
segments
and where
AR(*)
to
(221
< 0,5
avoid
interpolation
(j+m)-th
term
appro-
(16)
all
.
I
time
at
into
interval.
time
step
sampled
tj
current
i
,i,
Ri$s+l by,
:At/2
may
be
values
up
2
(sl;,t?] dt” = J!l
For reasoning
(23)
Pis
easily
the reflection
to be a constant
interval.
t”) .
TM
within
function
is con-
space
segment
each
This approximation
is satisfacto-
ry when u < Id” mho/m (fig. 2). On the contrary,
R;j-$($!t”)
will be interpolated
in time by a
Lagrangian
polynomial
,At/2
v+2
mZv R$j-s+l
TM
and NT
The
spacetime
then
be written
otherwise
t,$?m)dt!l
Ii+1 s+m
1
’
At/2
dependent
convolution
(24)
term
can
as
j+m +1
0,
) > 0,5
At/,
ti’ = to - tj)
of space
AR(*) = c(tR$’
for
include
Pijbi,tj)=Pijz
sidered
variation
(21)
[2].
problem
convolution
NT
jgl Iij (sY,tj’) u (s?) v(tjl)
NS is the number
the number
future
and time
s” zz s - s.
i
0
I’
with
for
want
[4]
repeated
bv :
= igl
we
to ti
of moments
spacetime
(20)
p.
1+1, jtm
I.
are the current
and convolui+l,j+m and Pi+l,j+m
rion term values at the center
of the (i+&tk space
TM
current
(191
1 I-1
the
(14)
moments
a
of a
in two dimensions
and
is .:
ds 0
0
function
Q,m!
B..
‘J
0
* -$-y(so,to*)}
(18)
(I,m) I,
i+l, j+m
“(9 =-I
because
Its integral
vc;j,
Bij
v’+2
JzV,
“(‘1
the
i and
where
+I v(l)+2 (so-si+ )(t(*)- t.+ )
(1 m)
p 0
B!$j
= II1
p=-1 pT$)(Si+l-si+p)(tj+m-tj+qJ
very
the low frequencies
we restrict
segment
response
isn’t
TM
It’s
. Method
V+2
rn%
and the
Gigg-0 (so,to)- K&o)$$g+o,to*)segment
the
convolution
and
the
Us;)
interpolation
pulse.
wire.
pij (sY,tT)
+1
=,5-l
pij(“,+
domain.
the
21
+1
= c
1=-l
MHz
dimensions
it
of space
interpolation
Lagrangian
Iij(s;,t;l
Fresnel’s
[9].
by
in neglecting
G
P(So,to*)
filter
lengths
II z) in the form
we can put Iij(sy,ti’‘f and pij(si,tj
in taking
finite
otherwire
t; = t* - t.
I
0
1
. Lagrange
be
SOLUTION
so like an infinite
cO
with
< _nz’
At the
= i;;
to
in time
Therefore,
simplicity,
wire
than
uncorrectly
III - NUMERICAL
J
intervals.
with
electroma-
using
generally,
to treat
:.Rs,t,=g
Ai and
time
mho/m,the
higher
pass
moments.
a long
with
method
electro-
But
method
obstacle
of the electromagnetic
to
of
(13).
inapplicable
interested,
the
verify
correctly.
approximation
in
frequencies
frequencies
wave
treated
one
the
coefficients
the
spectre
for
h = 10 m, E = 10, d = lo-’
r
condition
shows
that
only
frequency
gnetic
necessary
0,
i
p(so,to*)
be
It;1
Similarlv.
(13)
E
r
1,
I
=
to deduce
- 3,24 102’ (5’
would
for example
-
vet!;,
. Time domain
It
286
V+2
sZ1 r=C-1 t&l
.(r,t)
i+,
R$Mj+m-s+l
I~+,+~ s+t
,
(25)
287
where
v’ has been
taining
and
terms
used
later
in place
than
v
of
to avoid
5315
-
radiation
ob-
time t.
v-t2
in pulse
[l] JECKO
(26)
B. “Diffraction
sionnelles
developments
and
(18), (23) into
(14) allows
a
currents
current
matrix
at
convolution
the
integral
expression
time-step
v
to
for
be
equation
the
unknow
obtained.
It
is
(27)
Ziu is
the
between
v = 1, NT
u = I, NS
SELDEN
in
the
matrix
the
of
the
structure
mutual
interactions
segments.
It
is
space”
0.
It
Ii v are
specified
c&entvalues
a
repeated
to
sequence
bJ’
obtain
of
$1,
Ii2,
matrix
etc...
a
with
in
method
long
frequency
domain
frequency
[lo].
The
by lines
approximation
the
are
using
solution
the
4)
problem
inverse
validated
finite
which
differs
work
an
results
method
rencies
The
the
time
wire
is
is
obtained
evolution
h, the conductivity
tivity
greater
the
current
ted
its
is (Figure
wave
which
intensity,
6 and
the
the
the
current
different
height
later
is,
the
stronger
the
reflec-
the
diminution
and
later.
conductivity
and
of
Figures
permittivity
V - CONCLUSION
proach,
of the
by wire
of
a
treated
work
directly
electromagnetic
structures
particular
study
[8].
in
waves
of any
homogeneous
developed
constitutes
about
This
an
in the
It
first
time
scattering
form,
ground.
method
the
space
ted
(1972).
de
“Field
the
JECKO
3eme
(France)
computation
Mac Millan
B. “Diffraction
filaires
Colloque
Company
en
june
1983.
F.
“These
de
d’1.E.M.
presence
National
Electromagnetique.
University
T.K.
sur
la
Tregastel
Doctorat
of
domain,
problem
presence
continues
infinite
wire,
is
applied
ap-
the
already
to
the
“Analysis
thin-wire
ground
antenna
planes”
M.
New Brunswick
3eme
june
values
influence.
This
half-
vol.50
Limoges
of
structures
[71 SARKER
3eme
o and the permit-
instigates
antennas
conductivity
Doctorat
R.F.
Z&me
Limoges
de
3eme
(France)
cycle”
arrays
SC.
G.
over
E.,
“These
de
of
“These
University
J.P.,
imperfect
Universite
of
Doctorat
de
Limoges
(France)
1984.
de
of
Internal
Doctorat
Limoges
1983.
[IO] BERENGER
orien-
(1971).
University
january,
cycle”
of arbitrarily
Canada
M.P.
[91 LEBORGNE
5). Effectively,
arrives
7 show
of
for
of the height
E. The
des
sol”
no 2-84,
a little.
observed
G.3.
(1983).
et
[81 GOURDY
diffe-
IV.2. Influence of principal parameters
on
du
BURKE
(1968).
0.
II61MAUMY
(Figure
treat
execute
The
Transform.
forthe
compared
which
and
Ann.
to be published.
applied
. it’s
wire
methods
the
is
mdtalliques”
of Physics
de
Method”
New-York
cycle”
several
Fourier
by
february
(France)
operations.
IV-I. Validation of the method
The
impul-
of wire
a
University
par
I.
IV - APPLICATIONS
about
J.
Compatibilite
is possible
E.M.
“Analysis
“These
by Moment
time
) . ..) v - 1.
j=l
by
currents
of Ezv and the known
in terms
earth.
A.J.,
of
Canadian
151 DAFIF
the
E.S.
141 HARRINGTON
independent.
Finally,
the
(1983).
POGGIO
presence
cycle”
x=I,...,v-I
n”5-6
E.K.,
and
n”8-83
i = 1, NS
t.38,
[2] MILLER
term
[3] DAFIF
with
of
d’ondes
par des obstacles
Tdlecomm.
of
presence
BIBLIOGRAPHY
C\l,m)
Substitution
in the
report.
de
(France)
-
288
-
3500
F
3000
2500
1500
t
t
1500
-3
506
mhoim
I
t,(ns)
01
0
-
I’.‘.
“‘.
50
Figure
XXX
coo
4-
Current
wire
158
evolution
on
the
-
(ct.)
I
e
I’.“”
180
10
20
30
40
50
.
60
e
.
1
70
00
(In)
Figure
5 - Height
influence
can
be
used
to verify
the
condition
02) in
time
domain.
Here,
our method
using
the Fresnel's
approximation
(-)
is compared
with
a rigorous
solution
(---)
f‘ourlded on
the
Huyghens's
principle
(81
Fresnel's
approximation
Lines
approximation
Finite
differencies
I(t)
I(t)
500E
2508 -
400E
///,,///
N/Y AH
2000 .
3000
1500 -
2006
100a -
500 *
100e
t(m)
a~.."'..."*.'."
0
-
Figure
56
6
-
100
Conductivity
E
150
influence
-
Figure
1
10
20
7
-
.
I
I
30
40
Permittivity
1
.,
50
I.
I
60
70
influence
.,
00
et(m)
-
NOISE SOURCES
289
5416
-
AND INTERFERENCE
VALUES
IN HIGH VOLTAGE SUBSTATIONS
H. RGhsIer
A. Strnad
Energie-Versorgung
Stuttgart,
Abstract - Noise sources in high voltage substations
may cause severe interference
problems in the secondary circuits.
The knowledge of the different
noise
effects
sources and an analysis of the interference
allows to estimate
the overvoltages
to be expected
and their frequency of occurence.
By a proper design
of both HV equipment and secondary wiring and by the
application
of voltage limiting devices the noise voltages can be limited to an acceptable level.
1. Introduction
The phenomena of electromagnetic
transients
affecting electric and electronic
devices has been known
since the beginning
of electrical
engineering.
The
installation
of sensitive electronic
equipment
in high
voltage (HV) plants and the problems associated have
led to an intensive work in the field of interference
on
an international
basis.
Concerning HV substations a lot
been made mostly examinating
occured. However, up to now
description
of the noise sources
values.
of investigations
have
interference
problems
there ist no detailed
and the interference
It is the aim of this study to describe the noise sources
and to analyse the interference
values with special
regard to their shape, the maximum of amplitude and
steepness and the impedance of the noise source.
2. Electromagnetic
The noise
substations
noise sources in HV substations
sources which have to be regarded
are described below:
in HV
- switching in primary circuits (i.e. on the HV level)
Switching of disconnectors
or circuit breakers is a
frequent source of noise in HV substations.
The guided
waves are transmitted
by the current transformer
(CT)
and the voltage transformer
(VT) to the measuring und
protection
circuits.
Current
flow on cable-screens,
produced by guided waves and magnetic
fields, and
currents fed into the earthing system through CT and
VT generate
common mode voltages which also influence the secondary circuits.
- atmospheric events
A lightning stroke generates
travelling
waves on the
HV line. These waves can be produced by a stroke to
the conductor, to the earth shield wire or the tower.
The shape of the travelling
waves depends on the
amplitude
and the shape of the lightning current. A
flash-over
of the insulation can be caused by a lightning stroke to the line or to conductors in the substation, or by insulator contamination.
Whatever is the
Schwaben
AG
West Germany
cause of the flash-over,
it will produce electromagnetic waves which affect the secondary circuits. The.
lightning current fed directly or via an arc into the
earthing system may result in high potential differences within the earthing system.
- earth faults
Earth faults caused by the described events, by switching overvoltages,
conductor galloping or faulty switching are to be regarded mainly with respect to the
electromagnetic
waves radiated.
- switching in secondary circuits
De-energizing
of inductive
loads generate
transient
high frequency overvoltages in secondary circuits.
- electrostatic
discharge
Electrostatic
charged persons cause a very steep current with a rise time of some nanoseconds
when
touching earthed equipment.
- radiotransmitters
(walkie talkies)
The high frequency
field generated
by radiotransmitters,
including
those which are used by maintenance
staff,
can influence
sensitive
electronic
equipment.
The described noise sources affect the secondary circuits. It has to be distinguished
between the interference by conductive
(direct), inductive
and capacitive coupling on the one hand (guided waves) and
interference
by radiated waves (interference
fields) on
the other hand. The influence of radiated waves on the
secondary circuits working as antenna gets important
for high frequency events in the MHz-range.
3. Interference
values in open air substations
and GIS
Table I shows the maximum interference
values measured at the secondary equipment of open air substations and GIS (gas-insulated
substation).
The figures
are valid for careful installation
regarding the described
measures,
however
without
application
of
voltage limiting devices.
The interference
values and their origin are dealt with
following.
3.1 Guided waves
3.1.1
Switching
in primary
circuits
The most frequent
event
producing interference
is,
the operation of disconnectors
in HV substations.
During one operation
interval
there are up to 100 discharges of which the most critical follow each other in
not more than 10 ms.
-
affected
circuit
290
protectioncircuit
-
controlcircuit
auxiliary
supply circuit
type of
interference
guided
wave
radiated
wave
guided
wave
radiated
wave
guided
wave
radiated
wave
dimensions
kV
mT
kV
mT
kV
mT
0.3
1
0.3
1
0.3
1
atmospheric event
2.5
1.5
2.5
1.5
2.5
1.5
switching
atmospheric
earth fault in primary
event
circuits
0.3
2
1
0.1
1.5
1 .o
Noise sources
E
earth fault
z
21
2
$
?i
frequency range
open
air
substation
0.05-10
kHz
0.024
MHz
MHz
__I
:: (11
0.3
1.5
1
0.1
1.5
1
0.3
1.5
1
0.1
1.5
1
CJ
GIS
0.24
MHz
0.2-100
0.05-10
0.024
MHz
kHz
Note: The electric field is not considered; due to the screening always existing there is no serious interference.
The
figures are valid for the common mode voltages. In measuring transformer circuits common mode voltages and transverse voltages are identical
Maximum interference
values to be expected in I-IV open air substations
and GIS
with a rated voltage of 123.....420 kV
Table I:
The situation
is shown in Fig. 1. If, e.g., an opened
circuit breaker is isolated by a disconnector
from the
plant alive reignition occures as soon as the restriking
voltage across the disconnector
contacts is exceeded.
The shape of the wave transmitted
into the secondary
circuits is characterized
by a steep front with a rise
time of 100 . . . 200 ns in open air substations
and
5 . . . 20 ns in GIS. The high frequencies
in the MHzrange are able to influence secondary equipment seriously or even to destroy it. The maximum interference depends on the first ignitions when closing and
the last ignitions when opening the disconnector,
i.e.
the interference
rises with the restrike voltage and
hence also with the rated voltage of the substation.
The measured frequency range of the described events
is 200 kHz . . . 100 MHz for open air substations
and
GIS.
primary circuit
secondary
circuit
voltage of the arrester and thereby on the steepness of
the incoming wave. In open air substations
often only
power transformers
are equipped with lightning arresters; that is, why the measuring transformers
(VT and
CT) in the line section of a substation can be stressed
by high overvoltages
compared to the rated voltage.
Moreover these high voltages may cause a flash-over
of the insulation.
When steep overvoltages
or the breakdown of the
insulation
is transmitted
by the measuring transformers high transients
may occur in the secondary
circuits.
The frequency and the amplitude of these events depend on
- the transient
reponse of the measuring
transformers
- the statistical
distribution
of the lightning parameters
- the lightning
protection
facilities
of the HV line
and the substation
- the exposure area of the HV line
- the isoceraunic level
The amplitude of transients in secondary circuits caused by lightning phenomena and their frequency can be
estimated (see Appendix).
3.1.2.2
U, striking voltage
u secondary
Fig. 1:
3.1.2
voltage
Voltages in primary and secondary circuits
when opening a disconnector
(schematic)
Atmospheric
events
Amplitude and steepness of lightning currents show a
statistical
distribution
[ 1,2] ; consequently
the overvoltages expected can only be determined
by statistical investigations.
Establishing
an acceptable
risk
will fix the measures to be applied to the secondary
circuits.
3.1.2.1
Lightning
stroke to the HV line
,Lightning overvoltages
are limited on the HV level by
lightning arresters. The maximum overvoltage influencing the secondary circuits depends on the threshold
Lightning
strokes
HV substations
to earthed
components
of
Lightning strokes to the structures or earth conductors
of the substation or to the earth conductor of the line
close the substation
produce high transient
potential
difference
in the earthing
system.
These potential
differences
cause currents over the screens of secondary cables and the cabinets and earth connections
of
secondary equipment. Concerning the secondary cables
these currents produce voltages whose amplitude depends on the construction
of the screen, the length of
the cable and the amplitude
of the current
in the
screen.
It is possible to estimate the transient voltages caused
by lightning phenomena for a certain substation by a
combination
of measuring technique and mathematical
methods [3]: When discharging
a charged overhead
line into the earthing system of a substation the discharge current and the voltages at selected points of
the secondary circuits can be measured. The knowledge of these signals allows to compute the transient
response of the systetns. The frequency of lightning
strokes to earthed components
can be determined
by
-
291
5416
‘-
z
considering
the actual exposure
area of the substation
and the isokeraunic
level.
The maximum
transient
voltages
to be expected
at the selected
measuring
points
can then be calculated
by bringing
in the
statistical
distribution
of the lightning
parameters.
Fig. 2 shows the result of such an investigation;
for
the measuring
point M 3 the voltage
of 2700 V is
exceeded
once in ten years,
the corresponding
value
for M 4 is 560 V.
K
TK
Fig. 4:
1
Interference
by guided
waves.
H
t
The impedance
3.1.4
of the noise
source
10-l
The energy
content
of a noise voltage
is of great
importance
and has to be considered
especially
when
electronic
components
are stressed.
Consequently
for
a certain
shape of the noise voltage
the knowledge
of
the source
impedance
is necessary.
For a generator
constructed
to test the interference
withstand
capability of secondary
equipment
the proper simulation
of
the source impedance
is also an inevitable
demand.
_j_
a
10-a
IO2
Fig. 2:
103
V
T----)
“2
IO4
A model of an interference
in Fig. 4. Here are
Noise voltages
ii2 in a 420/123 kV open air
substation
caused
by lighting
strokes
to
earthed
components
H frequency
of strokes
producing
a noise
voltage which exceeds b,
M3, M4 measuring
points
the impedance
%Q
the voltage
u1
z,
Switching
in secondary
circuits
Overvoltages
produced
by de-energizing
inductive
loads occur because
of reignitions
during opening the
relay contact.
These overvoltages
are typically
sawtooth
shaped
followed
by a damped
low frequency
oscillation.
The amplitudes
of the overvoltages
can
exceed
5 kV, the risetime
of the spikes being in the
order
of some nanoseconds.
The frequency
of the
damped
oscillations
is normally
below
I MHz, but
sometimes
comes
up to 20 MHz. Figure 3 shows an
oscillogram
of the transient
voltage
generated
by
de-energizing
a relay coil. The application
of voltage
limiting
devices
allows to reduce
the transient
voltages to less than 1000 V.
impedance
time
the frequency
the secondary
dependent
equipment
of the secondary
of the secondary
input
cable
impedance
at the end of the secondary
the resulting
-za
is shown
of the noise source
the travelling
the voltage
u2
waves
of the noise source
the characteristic
cable
ZK
‘K
3.1.3
by guided
impedance
of the noise
of
cable
source.
The characteristic
impedances
of secondary
cables
(conductor-conductor;
conductor-screen)
come up to
30 ..* 100 ohms nearly independent
of frequency;
however, these figures are not valid for the circuit screenearth return. In most applications
we have zK L Z+.
During
Z+
2
TK
= zSQ;
we
then
After
3 T,
approximation
for
ist charged
2
T~L
t L
4 TK
to
+ ZSQI.
by the
an alternating
Z+ = z,,
the cable
u2 = 2 UIZK/GK
Caused
find
reflections
behaviour,
with
at the cable
the final value
T = z,,
to zsQ
’ CK
and
ends Z_ shows
being
CK =
2
-.SQ’
T~/Z~,
the
is sufficient.
U2
1
J
--Y-
200 V
3.2
+I I-
3.2.1
2 ms
Fig. 3:
Transient
voltage
zing a relay coil.
During t L 2 T the characteristic
impedance
of the
secondary
cab lK
e represents
the impedance
of the noise
source.
In HV substations
the length
of secondary
cables is about 30 m to 100 m; assuming
a travelling
wave
velocity
of
0.16 m/ns
we
find
for
t = 0.375 US . . . 7.5 ,.us Z. = ZK. In most
cases
the
I
maximum
of the noise voP tage- will appear within this
period.
generated
by de-energi-
Radiated
waves
Switching
in primary
circuits
The magnetic
flux density
produced
by switching
in
420 kV substations
comes up to 0.1 mT nearby secondary
cables
a
range
of
in
frequency
-
200 kHz . . . 100 MHz. For the
10 kV/m have been measured.
electric
field
up
292
to
In open air substations
e.g. the disconnector
switches
the current transformer
and the line section between
disconnector
and circuit breaker. When neglecting the
resistances one finds fo: thf current from CT to earth
with C as the
caused by discharges
I = uLE /m
conductor-earth
capacitance
of the CT and L as the
inductance
of the circuit. For 420 kV substations
it
can be estimated
C = 1 nF, L = 50 uH . . . 100 ,uH resulting in “; = 1085 A . . . 1534 A. Thi.$ current produces
in a distance of 5 m a maximum magnetic flux density
B = ‘i.p,/2n r 1 61 ,uT.
When operating a discon;ecFr
in GIS the correspon, = uLE/ZLd = 1372 A
with
is
current
ding
2 = 250 ohms as the source lmpe ante of a 420 kV
okerhead
line. This current
will flow at the GIS
bushing along the enclosure and via the earth wires to
earth.
3.2.2
Lightning strokes
substations
to earthed
components
-
By a proper choice of the material (wool, antistatical
treatment)
and/or regulation of the humidity the above mentioned value can be met.
3.4 The shape of the noise voltage
Transient
voltages
in HV substations
are normally
strongly damped. To simulate
a noise voltage by a
damped oszillation according to[ 71 will cover a lot of
the interference
occuring. However, the rise time of
of
75 ns fixed in [7] d oes not meet the requirements
GIS; here rise times of 5.....10 ns have to be considered.
The amplitude of switching overvoltages
varies during
the operation interval. Secondary equipment may show
maloperation
when voltage limiting
devices are applied and the transient
voltage amplitude
is right
below the limiting voltage.
4. Measures to reduce interference
HV substations
values in
in HV
4.1 Primary
circuits
Lightning strokes to earth shield wires and structures
will normally lead to a distribution
of the lightning
current which is fed via a number of injection points
into the earthing system. However, if lightning rods
are installed in form of steel masts erected on the
ground the current is injected at one point into the
earthing system.
The maximum magnetic
flux densities
can be estimated as follows:
There are only a few steps possible to reduce the noise
sources in primary circuits. This stems from the fact
that there are few technical
solutions which can be
implemented
at reasonable
cost. Therefore
most of
the measures listed below are necessary to protect HV
equipment,
but they have also a beneficial
effect on
interference
in secondary circuits.
From the statistical
distribution
a lightning current
amplitude
of 200 kA can be withdrawn
as a 95 %
value, i.e. in 5 % of all events this value is exceeded 3 . Concerning 4 injection points into the earthing
system the magnetic flux density in a distance of 10 m
is B = 1 mT. For only one injection point the distance
between the lightning rod and the relay kiosk may be
about 15 m; in this case the maximum flux density is
B = 2.65 mT. The lightning
current
resp. the transmitted field has a frequency range up to some MHz.
Nearby HV substations,
especially
nearby GIS, overhead lines are equipped with two or even three earth
shield wires. This measure reduces the frequency and
the amplitude of strokes to the conductor. By this the
influence of lightning strokes on secondary circuits is
reduced, too.
3.2.3
Earth faults
The highest 50 Hz flux densities are to be expected for
earth fault currents. In a 420 kV substation for a fault
current
of 35 kA the magnetic
flux density
is
B = 0.1 mT considering a distance of 7 m.
3.2.4
Radiotransmitters
(walkie talkies)
The intensity of interference
fields produced by radiotransmitters
depends on the distance between antenna
and secondary equipment and on the transmitted
power. An acceptable
maximum
value seems to be
10 V/m [4,5].
4.1.1
4.1.2
3.3 Electrostatic
discharges
The electrostatic
charge can be limited to a value of
5 kV also for substations.
The voltage people can be
charged up to depends on the material of the carpets
in control rooms and on the relative
humidity; for
lower humidity the maximum voltages rise strongly 161.
against
Configuration
lightning
of earthing
strokes.
system
An earthing system properly designed for 50 Hz stresses needs only slight modifications
to reduce transient
voltages. Firstly single tee off connections
have to be
avoided in general,
and secondly
bare conductors
should be layed parallel to long cables and be connected to the earthing grid.
4.1.3
Measures at CT and VT
The transient
voltages transmitted
via the measuring
transformers
to the secondary circuits can be reduced
by careful earth connections
within the transformer
and additional screening of the secondary windings 191.
4.2 Secondary
4.2.1
The permissible distance d depends on the transmitted
power P and is d = 1.6*/-P/E [6]; for E = 10 V/m and
P = 2 W the permissible distance between antenna and
cabinet is d = 23 cm.
Protection
circuits
Screening
of secondary
cables
The best system to reduce interference
in the circuits
is the adoption of screened cables. In the ideal case of
continuous and perfectly homogeneous screens with no
resistance,
the protection
against the external
high
frequency
electric and magnetic field would be perfect. Because of the practical
performance
of the
cable screen however, one has to consider some points:
-
The screen should be almost continuous and with
low resistance (a few ohm/km).
The screen should have a low coupling impedance
within the interference
frequency range.
‘-’
-
_
Earthing of the screen should have a very low
impedance, that is, the earthing conductors should
have adequate
section,
minimum
length an optimum contact arrangements.
In some cases it may be necessary to earth the
screens at the inlets to the relay rooms or in the
equipment cabinets so that the currents circulating
in the screens do not affect the unscreened
circuits; whenever interferences
are due to induction,
earthing at both ends is suitable.
4.2.2
Configuration
of secondary
circuits
The cable route should run as far as possible
not parallel to bus-bars or power cables.
-
293
from and
The forward and return conductors
of the same
circuit should be run in the same cable; twisted
pairs or quadruple cables should be adopted whenever possible (for instance
for very low current
circuits and data lines).
5416
-
4.3 Introduction
of new technologies
An essential reduction of interference
can be achieved in the future by
-
by guided waves
application of optic fibre cables
development
of electronical
measuring
tratX.fOrmerS
transmitting
the digitized
signal by fibre
optic cable from the HV level to the protection
equipment
It is the advantage
neither conductive,
nor radiated waves
sion. Consequently
mary events will be
of the fibre optic cables that
inductive and capactive
coupling
are able to disturb the transmisthe noise voltages caused by prineglectable.
Today, CTs or VTs with digital output are still in the
experimental
stage and their introduction
on a wide-spread basis is not expected in the near future.
5. Conclusions
-
All the screened
cables should run ,as close together as possible in order to benefit from their
mutual screening effect.
-
Laying of bare conductors
parallel ,to the cables;
the conductors are to be connected to the earthing
network at the two ends and, if possible, at a few
points along their route.
-
for D.C. auxiliary
supply cables
guration is better than a ring.
4.2.3
Use of voltage limiting
a radial
confi-
devices
Voltage limiting devices should be installed inside the
protection
and control equipment;
their installation
outside the equipment in new plants should be avoided.
However, their adoption might be useful in plants
already in operation, to allow installation
of standard
devices with low EMC limits.
The most common
-
-
-
-
voltage limiting
A proper and careful installation
leads to interference
values in secondary circuits of HV substations
which
are below 1000 V; an exception are the noise voltages
caused by lightning strokes and by switching in secondary circuits. As the lightning current amplitude and
steepness
is statistically
distributed
the acceptable
risk determines
the measures
to be taken. Ry installation of voltage limiting devices the amplitude of
the noise voltage can be held within the acceptable
limits. The same concerns the noise voltages produced
by secondary switching.
The future use of optic fibre cables will result in an
essential reduction of interference.
Future measuring
transformers,
equipped with digital electronics
and
fibre optics will transmit only small noise voltage into
secondary circuits.
devices are:
Condensers
or RC circuits that reduce HF overvoltages and are adequate both for circuits coming
from the switchyard
and for supply circuits.
An
interesting
application
consists
of filtering
the
auxiliary D.C. power of circuits and equipments at
the lowest EMC level, supplied by a battery which
is feeding also circuits
and equipment
(e.g., HV
breakers and disconnectors)
at the highest EMC
level.
LV arresters discharge overvoltages
at high energy
content
in A.C. and D.C. circuits
having voltages -5 48 V and have such a time delay that they
are not adequate for steep and HF overvoltages.
They require
a low-impedance
earthing
to the
equipment
they protect;
generally
they are installed on telecommunication
lines.
Zener diodes are advisable for overvoltages
at low
energy content only and therefore
should be used
with great care. Transzorb diodes are electronic
components similiar to zener diodes with very good
characteristics
such as very small delay time and
leakage current and constant voltage limitation
up
to some hundreds of volts. With a series-parallel
combination
they can be used for high voltage
limitations and for high energy transients.
Varistors show a restistance
that is inversely proportional to the applied voltage; utilization
of the
zinc-oxide type is spreading (there is a tendency to
replace zenerdiodes
and RC circuits
with these
varistors) because of their short delay time (5 25 ns)
and high impulse current (up to 25 kA).
Appendix
Estimation
of transient
secondary
lightning strokes to the conductor.
voltages
caused
by
a) 420 kV open air substation
For a shielding angle of 45” (I earth shield wire) the
maximum lightning
current
amplitude
reaching
the
conductor
is I
- 36 kA [I]; the measured transient
secondary
vol aie caused by switching
in primary
circuits (U = i ‘-I-l /p) comes up to U = 500 V.To
estimate
h&w oftenmU = 2500 V is excee ?led at first
the corresponding prim&y voltage Ul is determined:
resulting
2.1710
‘Bl=
kV
=
13.7
in
kA
ZW
of the
with 2 = 250 ohms as the surge impedance
overh&x line. Lightning currents between 13.7 kA and
36 kA can produce secondary transients
of 2500 V and
more.
It is known from measures that the steepness S of the
primary voltage ist proportional
to the amplitude of
the secondary transient
for S 2 10 kA/,us. Since 97 %
of all strokes have a steepness 2 10 kA/ us anyhow this
value is fixed as a bottom limit fo t/ the steepness
considered.
234
125
T;:
i
::
1
100
!
!
/
Fig.Bl
50
25
25
0
50
0
100
150
200
250
3;0
I
L=
0.8
1
1
kA/ks-z
w/km
1
= 3.81
km
For this value of L the maximum steepness stressing
the substation
is 10 kA/ us. Along the collection
length L the steepness of t h e lightning
.
strokes causing
10 kA/ us at the substation
will differ; the relations
are sh d wn in table A for different length 1. P describes
the probability
that the corresponding
steepness
ist
exceeded
for stroke currents
from 13.7 .. . . . 36 kA.
These figures can be derived from the presentation
in
fig. 6 showing a typical distribution
of amplitude and
steepness of stroke currents
when considering
their
correlation 131.
The integral of the probability P along the collection
length L comes up to P, = 24 %, i.e. along L 24 % of
all strokes
fulfil
13.7 kAL IB” 36 kA and cause
S b 10 kA/,us at the substation.
The frequency H of
secondary transients
1 2 500 V is then
H = B-L. P;k.NB
with
collect.ion width
LB isokeraunic level
N”, z “1!9;a
km21
[l]
k regards which part of the strokes within the determined collection
area really hit the conductor.
Following GOLDE [lo3 it can be estimated k = 0.25.
H = 0.04 ’ 3.81*0.24.0.25
= 0.017/(a.circuit)
- 1.9/(a=circuit)
This results in a transient secondary voltage h 2 500 V
every 58 years for each circuit; a substation with five
circuits will be stressed every 12 years with such a
transient.
Table A
S (kA/,us) 65
60
50
40
35
30
25
20
15
10
l(km)
3.81 3.75 3.60 3.38 3.21 3.00 2.70 2.25 1.50 0.00
PI%)
0.0
0.2
0.8
2.4
4.6
8.0
12.6 23.6 32.6 39.2
40
35
30
25
20
15
Table B
S(kA/,us)
43
I(kd
1.53 1.50 1.43 1.33 1.20 1.00 0.67 0.0
PC%)
0
0
0.2
1.0
1.6
4.2
10
10.2 14.4
0
50
100
150
200
250
I
[LA]
From Fig. A the 95 % value of the steepness corresponding to IB2 = 36 kA can be derived 5 = 65 kA/ US
[3]. Regarding an attenuation
of 0.8 ps/km along t!he
line the collection length L becomes
IB2 * 10
0
300
bA1
b) 420 kV GIS
Considering
two earth shield wires the maximum
- 16 kA, the
stroke current to the conductor
is I
95 % value of the steepness is 43 kAv2ui. The measured transient secondary voltage cause 6 by switching
in primary circuits is U2 = 1000 V. To estimate
how
often a secondary voltage of 2000 V is exceeded the
corresponding
primary voltage U - 684 kV and the
stroke current j-,1 = 5.5 kA have take considered.
The collectlon
ength 1s L = 1.53 km, from table B one
gets pZ= 7 % and the frequency of transient
voltages
exceeding 2000 V is
H = B*L* p,- k*NB
= 0.025 * 1.53. 0.07 * 0.25. 1.9/(a.circuit)
ZZ0.0013/(a.circuit)
i.e. a substation
with five circuits will be stressed
every 157 years with a secondary transient A 2000 V.
References
PI Anderson,
meters for
R.B.; Eriksson, A.J.: Lightning
paraengineering application.
Electra Nr. 96,
1980, 65-102
von
PI RGhsler, H.; Strnad, A.: uberspannungsschutz
metallgekapselten
gasisolierten
Schaltanlagen
im
420-kV-Netz.
Etz Archiv 6 (1984), 233-238
PI Fischer, M.; Strnad, A.: Bestimmungen der bei
Blitzeinschlsgen
zu erwartenden
transienten
Uberspannung in Sekundgrkreisen
von Hochspannungsschaltanlagen.
ElektrititZtswirtschaft
82
(1983)
87-91
P+JAnders, R.; Campling, A.C.: Interference problems
on electronic
control equipment
in power plants
and
substations - installation
and interference
tests. CIGRE-Report
36-05 (1980)
G.: Disturbances
produced by transr51 Champiot,
ceivers and walkie-talkies.
Electra No. 83 (1982),
pp. 103-110
conditions - Part 5: Electro[61 IEC 654-5: Operating
magnetic compatibility.
Draft-Publication
1983
Impulse voltage withstand
tests and
PI IEC 255-4:
high-frequency
disturbance tests. Appendix E.
PI Riihsler, H.; Strnad, A.: Die HBufigkeit riickw;irtiger Uberschkige
und ihre Reduzierung.
Elektrizit8tswirtschaft
82 (1983), 386-390
von Sekundsrkreisen
in
PI Strnad, A.: Beeinflussung
Hochspannungsschaltanlagen
bei rasch ver;inderlichen VorgZngen im Hochspannungskreis;
Thesis
(1982) Technical
University
of Darmstadt,
West
. Germany
[IOj Golde, R.H.: Lightning,
Vol I. Academic
Press,
London 1977
-
BALLOON
AND SATELLITE
295
-
55
OBSERVATION
OVER NORTHERN
Takeo Yoshino
University
l-5-1 Chofugaoka,
17
OF POWER LINE RADIATION
EUROPE
and Ichiro Tomizawa
of Electra-Communications
Chofu-shi,
Observation of the electromagnetic
field variation phenomena of Power Line
Radiation (PLR) related with polar substorm activity had been done by two
observation balloons and one satellite
in the arctic circle of northern europe.
The balloons named as B15-1N and B15-2N
were launched on March 20, 1982 from
Stamsund, Norway and on November 23,
1982 from ESRANGE, Sweden respectively.
Both balloons could be obtained several
electromagnetic
field variation data
with various frequencies caused by substorm effect during their flight.
By the balloon observation, the
field intensity of higher harmonic
frequency of Power Line Radiation(PLHR)
in the frequency range between 200Hz to
1 kHz which were enhanced by the EM
field disturbance in the poler substorm,
were obtained the clearly spike-like
level increase as harmonics of 50 Hz
step in the frequency spectrum data, at
300 Hz, 450 Hz and 600 Hz appeared in
the data of B15-1N and at 300 Hz and
450 Hz in B15-2N. However, the intensity increase of fundamental, and 2nd
and 3rd harmonics of 50 Hz were not
exceed over than 3 dB during the polar
substorm.
The Japanese scientific satellite
EXOS-C "OHZORA" was launched on 14th
February 1984. The data in the northern
europe is received from satellite at
ESRANGE station based on the JapanSweden co-operation program for EXOS-C,
and the data with analized form will be
able to appear by end of this year.
INTRODUCTION
As shown in a recent paper [l], the
observation results of fundamental ELF
wave propagation characteristics
of
Power Line Radiation to horizontal and
to vertical direction were observed by
using of balloons and rockets over
-Japanese island. One of the purpose of
this experiments was the determination
of the standard propagation characteristics of ELF waves during the condi-
Tokyo
182, Japan
tions in quiet solar activity at middle
latitude area used as the caribration
standard. By this observation resultsI
the attenuation constant to horizontal
direction propagation which are consisted of a guided mode between the
bottom side of ionosphere and the
ground (sea) surface observed by balloon experiments, is approximately
1.2 dB/lOO km at 50 Hz and 1.3 dB/lOOkm
To vertical
at 60 Hz respectively.
direction propaqation which penetrate
into the lower ionosphere is-approximatelv 1.1 dB to 1.2 dB/lOO km on
eithe; frequency, which'observed
by
four rocket
experiments as described
in the recent paper [l].
This work continue after publish of
paper [l] to obtain more precisechracteristics on the several items of
basical propagation characteristics
at
more wider areas. For example, a transPacific balloon experiment have been
launched on September 23, 1984 from
Sanriku test range, Japan, to observe
the long distance propagation over the
Pacific ocean of Power Line Radiation
from Japan. And this balloon flew eastwards and operated 50 hours after
launch, and the flight distance was
approximately
2000 km on the pacific
ocean from Sanriku, Japan.
The 9th Japanese scientficsatellite
named EXOS-C "OHZORA" was launched on
3.4th February 1984, from Kagoshima
Space Center as anouncement in the
paper 111 for global monitoring of PLR.
The observation results of this satellite will be mentioned later.
In the last decade, lot of high
power hydro-electric
power generator
stations have been built in sub-polar
regions or in polar regions as Manitoba
and Quebec in Canada and in northern
area of Norway and Sweden. And these
huge electric power have been carried
by extreme long distance transmission
line with 3 wire system over 2000 km
from sub-polar area to main industrial
area in the middle latitude,across
perpendicular the aurora1 oval. The
line voltage of these power lines are
-
296
This paper presents the results of
balloon observations on the electromagnetic field induced by Power Line Radiation observed around aurora1 oval
under geomagnetic 'active conditions.
Observations have been done by using
balloons, that is why, strong electromagnetic radiation of Power Line field
will be suppress the true values of
integrated electromagnetic
field
induced by power line systems.
500 kV or more.
At major geomagnetic disturbavce on
September 21, 1977, the huge unbalance
anomaly current of over 25A to 50A was
induced, and observed this induced
current at the grounding point of the
transformers on the Manitoba line of
Canada. And there are many troubles
appear on this line such as the breakdown of main circuit breakers and the
burnout of transformers wires. The
many similar troubles in this time
appeared not only on the power lines
of the Manitoba lines but also on the
other long power line system at subpolar and polar area as Alaska, north
Canada and northern europian areas.
At same time, the troubles were expanded other field as long distance
communications
wire and the pipe lines
in the sub-polar and polar regions. [2],
[3] and [4].
Above mentioned induced anomaly
current on power lines induce magnetically saturation of transformers core,
and the waveform of current and voltage
will be distorted by the non-linearity
effect of the saturated transformers.
This distorted current will be produced
lot of higher harmonic frequencies
radiation from power line system to
magnetosphere. [5,6].
BL5-2N
-
OBSERVATION
BY BALLOON
The balloon B15-1N was launched at
1909 UT on March 20, 1982 from Stamsund
in Norway, and the balloon B15-2N was
launched at 2059 UT on November 23,1982
from ESRANGE in Sweden. The flight trajectries and altitudes of balloons are
illustrated in Fig. 1. The trace of
B15-1N drifted eastwards from Stamsund
and flight to across over northern
Norway and Sweden, and dropped the payload in the territories of Finland as
shown in Fig. 1. And B15-2N drifted
toward north-east from ESRANGE, and
flying across over northern Finland,the
west of Kolskiy area of U.S.S.R.and the
payload dropped into Arctic ocean at
off the coast of Murmansk as shown in
Fig. 1.
(1982/11/23-24)
70
(1982/ 3/20
U.S.S.R
1lJ
1982/
3/20
.I
0
15
20
25
30
35
40
45
50
55
6
DE
LONGITUDE
Fig. 1.
The flight trajectries and altitudes of balloons Bl5-1N and B15-2N. The
balloon B15-1N was launched at 1907 UT on March 20 and the balloon B15-2N
was launched at 2059 UT on November 23, 1982 respectively.
-
297
5517
-
ELECTRIC
FIELD
,
PRE
AMP
AMP
E
PRE
AMP
AMP
HPF
U-MAGNETIC
FIELD
I
I
1
L
Fig.
2.
A schematic
block diagram
line radiation measurement
board the balloon.
of the power
system on
The geomagnetic activity on flight time
of both balloons, Kp index was 2+ from
19 UT to 21 UT on March 20, 1982 for
B15-lN, and the value of Kp index
fluctuate from 6+ to 3-l-from 21 UT on
November 23 till 03 UT on 24, 1982 for
B15-2N.
The on boarded equipments of balloons B15-lN.and 2N have mostly same
properties except the receiving gain.
Fig. 2 shows a block diagram of PLR
detector, and the frequency range of
this detector covers from 40 Hz to
1 kHz. The detector consists of a horizontal magnetic field sensor, a vertical electric field sensor, and a multiplexer for switching the magnetic and
the electric field signals as shown in
Fig. 2. The detected signals are
connected to the telemetry transmitter
through IRIG-#18 FM channel, which has
the maximum frequency responce of 1kHz.
The horizontal magnetic field
sensor is equipped with a loop antenna
which uses a permalloy core (50cm x 6
mm x 6mm) wound 6000 turns with the
copper wire of 0.2 mm pl, a flat frequency responce pre-amplifier,
and 40 Hz
HPF which is required to reject the
strong Schumann resonance frequency
portion on the spectrum.
The vertical
electric field sensor is equipped with
a short electric dipole antenna which
has a copper wire of 10 meter tip-totip, connected with high input impedance terminal of pre-amplifier,
and
1 kHz LPF.
The multiplexed signal is
separated into two amplifiers in which
the gain difference is 40 dB to get
wider dynamic range.
OF OBSERVATION
RESULTS OF Bl5-1N
Fig. 3 (a) shows the H component
variation of geomagnetic disturbance
at Andoya during flight the B15-1N
balloon.
The Figures 2 (b) and (c)
g
I
l-i
;;
2
:::
i 6)v-i
g
5
E
5
i
_
‘p
4
ml9
s
;
;
';
6
i
z
L
!:
g
d :
i !2UT
21
20
1982/
3/20
5OHz
I
,60HZ
\lBOHz
ml9
20
21
.-I
Fiq.
RESULTS
are display of the observation
record of on board detecter on
the variation of magnetic and
electric field intensities respThe contours of 60 Hz
ectively.
and 180 Hz as indicated by dotted
line are shown as refer to the
But
back ground noise level.
the real electric field intensity
does not exceed over the system
noise level in this figure for
the gain set too low against the
intensity of back ground noise
level in this district.
In the Fiq. 3 (b), the magne-,
z2UT
7JME
1982/ 3/20
3.
(a) The horizontal
component of
the masnetograph at Andoya, Norwayl
(1;) and (c)the maanetic and
electric field intensity which
observed by balloon B15-1N on
March 20 1982. Solid line indicated 50 and its harmonics, and
dotted 1 i ne indicated 60 and 180
HZ.
I
-
298
tic field intensity at 50 Hz has a pea %
at 1936 UT with the value of 1.6 x lo-,
greater than the
(A/m) which is much
,
_
value of 60 Hz (rerer to back ground
The intenoise level close to 50 Hz).
nsity at 50 Hz gradually decreases with
time until the end of data without any
enhancement, even at the time of substorm onset at 2040 UT. The intensities
of harmonic radiation at 100 and 150Hz
are less than the intensity of background noise through out this observation.
The magnetic data between 1909UT
to 1920UT should be omitted, because
the large swing of contours are affected by the joggle of gondola during the
ascending of balloon.
Dynamic spectra of the magnetic field in the frequency ranqe between 40Hz
to 1kHz is shown in Fig; 4.
Spectral peak of 50Hz is
clearly identified until 20
13UT, however, after 2017UT
the 50Hz sunk into background noise level. Spectral
peak at harmonics of 50Hz
cannot be identified in the
dynamic spectra until 2036UT.
At 2040UT, a spectral peak
appears at 350Hz, but it
does appear after 2044 UT.
The remarkable spectral
peak appears in the successive spectra between 2044UT
until 2121 UT. Also the
spectral peak at 600 Hz,
peaks at 300 'Hz and 450 Hz
appear in the spectra from
2051 UT to 2121 UT, but the
peaks of 300 and 450 Hz are
not clearly identified. It
is noticeable that the peak
frequencies are appeared
not exactly at the harmonic
frequency of 50Hz, occasionally shifted lowerside.
These lowerside shift of
frequencies of the three
spectral peaks is correlated
each other, and the amount
of shift frequency is proportioal to the frequencies
of 300, 450 and 600 Hz. This
lowerside frequency shift
bill be attributed to the
shift of fundamental frequency of 50 Hz. The appearance of these spectral peaks
is coincidence with the geomagnetic substorm which
started at 2040 UT. Therefore, it is concluded that
the geomagnetic substorm
not only enhances the power
line radiation especially at
300,450 and 600 Hz, but also
disturbs the frequency of
the generator revolution as
0
heavy unbalance load.
Fig.4 Dynamic spectrum of
the magnetic field in the frequency
B15-1N balloon.
REXULTS OF B15-2N
Fig. 5(a) is shown the magnetgram
record of H-component magnetic field
intensity and ULF magnetic pulsation
on November 23, 1982 which observed at
ESBANGE, Sweden, during the flight of
Bl5-2N balloon. A polar substorm are
occured at 2251 UT and decayed after 30
minutes. A negative bay of H-component
of magnetic field and a sudden comencement of H-component of magnetic pulsation at this substorm were recorded in
Fig. 5(b) and 5(c) illustthis figure.
rate the magnetic and electric field
intensity variations during the balloon
and
flight at 50Hz, 60Hz,100Hz,150Hz
180Hz respectively. The contours of 60
Hz and 180 Hz indicate of background
noise level close to the fundamental
0.5
Freqency
range between
LkHzl
40Hz to 1 kHz observed
by
-
299
and 3rd harmonics of power line radiation. The magnetic field data between
2115 to 2220 UT should be omitted,
because the large swing of ccpztaurs
are
affected by the joggle of gondola
during the ascending of balloon. It
gives a clear evidence that the intensities at 50Hz,lOOHz and 150Hz were
not enhanced during substorm.
The electric field intensities of
50Hz, 1OOHz and 150Hz had increased
from 0115 UT to 0230 UT in spite of
low geomagnetic activity, when the
balloon flew over the large industrial
and mining area near Mulmansk in USSR.
The level of enhancement of the magnetic component are much less than the
electric component at this time. This
record is suggested that the enhancement of power line radiations could be
induced by a localized power line
system. The Lapland area have not much
high voltage power lines. Fig. 6 shows
the dynamic frequencyspectra of electric field intensities at this time.
r
I
FREQUENCY (kHz)
Fig. 6
I
I
t
21
22
23
1
I
I
00
I
I
‘
01
02
03
-4
7
6X21
I
22
23
00
01
I
.
02
I03
Dynamic frequency spectrum of
electric field intensities at
0115UT to 0230UT by balloon
Bl5-2N.
Fig. 7(a) and 7(b) shows the amplitude spectra of magnetic and electric
fields around 2250 UT of the frequency
range between 4OHz to 1kHz of PLR
harmonics radiation during the geomagnetic substorm which started at 2251UT
as shown in Fig. 5(a).
From the observation results in
this figure, it is clear evidence that
the spectral peaks at the harmonic frequencies of the power line radiation in
the frequency range higher than 200 Hz
are enhanced at the time of the substorm onset at 2251 UT in the electric
and magnetic fields. The intensity in
enhancement at 300 Hz and 450 Hz corresponds to the duration of the enhancement of ULF pulsation included high
frequency component as shown in the
Fig. 5(a).
OBSERVATION OF SATELLITE EXOS-C"OHZORA"
The Japanese scientific satellite
EXOS-C "OHZORA" was launched at 0800UT
on 14 February 1984 from Kagoshima
Space Center in Japan. Satellite on
boarded the observation equipments for
correpondence to MAP program and a 3
channel narrow band receiver for magnetic field detection and a wide band
2:
receiver for electric field detector to
cs21
22
23
00
01
02
03UT
PLR observations. The initial orbital ’
data was; apogee-865 km, perigee-354km,
1982/11/23-24
incrination-74.5'and orbital period-96.9
Fig. 5 Intensity of power line radiation
minutes. The data in the northern europe
observed on the balloon Bl5-.2N,
is received at ESRANGE station, Sweden.
(a) H-component of magnetgram and
The
data of PLR in northern europe is
ULF magnetic pulsation.(b) and (c) handling
for analysis, and the presenmagnetic and electric field inten- tation of this results will be able to
sities record at 50 to 180 Hz.
start by end of 1984. As an example, a
data observed over east China shows as
Figure. 8.
.-I
- 300 -
CONCLUSION
ill
References
Yoshino T,, and I.Tomizawa: Rocket &
balloon observations of power lines
over Japanese islands, EMC 81,Zurich,
pp525-530,(1981).
Boerner, W-M., Res. Grant Proposal
(1979 and 1980)
Akasofu, S-I., and Merritt, Nature,
279, 308-310,(1979)
Lanzerotti, L.J., Space Sci. Rev.,
34, 347-356, (1983).
Helliwell, R.A., et al, J. Geophys.
Res., 80, 4249-4258, (1975).
Luette, J.P.,,C:G. Park and R.A.
Helliwell, Geophys. Res. Let.,4,275278, (1977).
Hayashi, K., et al, Nature, 275,
By the balloon and satellite observations of power line radiation characteristics, the relationship between the
power line radiation and the magnetic
[21
substorm must be separate to estimate
at the effect of the frequency range
[31
below third harmonics, and at higher
than 4th harmonics of fundamental fre[41
quency of power line for three phased
AC transmmision system. The intensity
[51
of fundamental frequency and lower
harmonics of PLR observed on the ball[61
oon is not controlled by the geomagnetic substorm. And it is suggested that
the enhancement of power line radiation
[71
at lower frequencies will be influenced
627-629,
(1978).
mainly by the localized condition of
[81 Park, C.G.and R.A. Helliwell, Science
power line networks and power consumpScience, 200, 727-730, (1977).
tion in that area.
t91 Tomozawa, I. and T. Yoshino, Memoir
However, the intensities of the
of NIPR, Special Issue 31, 115-123,
higher harmonics more than 4th orders
(1984).
are enhanced by the magnetic field
variation of substorm as shown in the
Figures 4 and 8. This phenomena was
also observed clearly
on the ground at the
time of geomagnetic
b)
subsrorm, which increase
of 3rd, 6th, 9th, 12th
and more higher harmonics of power line. This
data appeared in Central
Canada in 1979 [7].
Ry the radiation
mechanisum described the
recent paper 111, the
intensities of radiated
magnetic field are emitted from a current loop
r!
which consists of neutQ,
ral wire of three phase
iT
system and the ground,
.umust be intensified by
these harmonics current. E
The intensity of PLHR at
g
2
more than higher frequencies can be even induced under the condition of the more weak
1.
a.*0 1
0.5
0.5
0.0
1.
FRE:QUENCY
substorm. [8,91.
tra
of
thdkE
&tric
_I
and
magnetic
Fig.
7
Dynamic
spec
The data of this
field betwee n 40Hz-1kHz observed by B15-2N.
balloon and satellite
I ’ ‘
experiments will be able
Rev
to offer the lot of
qls:
valuable saggessions on
the research field to
related effect of power
line radiation in the
magnetosphere.
T--l
l-u 1
Acknowledgement
The authors 'are deeply
indepted to Drs. H. Yarn;-gishi, T.Ono, H.Fukunishi,
H.Miyaoka and M.Ejiri of
NIPR, and the staff of
ESRANGE, and the staff of
ISAS to their cooperation
and to helpful discussion,
and to the assistance of
S. Yamakawa in the spectral analysis.
I
I
I
L
121 111 111 1s 181
-'&s' ’ ’ * ’ * * . .
LONGITUDE
1984/6/2 UT
Fig. 8. The RLR observation over eastern China by
satellite EXOS-C "OHZORA",
171
- 301
56~1
-
THE NUMERICAL ELECTROMAGNETICCODE (NEC)"
J.K.Breakall, G.J.Burke, E.K.Miller
Lawrence Livermore National Laboratory
Livermore, CA 95550
I.
INTRODUCTION
In the 20 years or so since the first
computer-based Moment-Method solutions for EM
radiation and scattering problems were
presented, numerous computer codes have been
developed. Relatively few of them have found
applications beyond those for which they were
originally intended. Of the latter, furthermore, only a small fraction have received
widespread acceptance and implementation.
The Numerical Electromagnetics Code--Method
of Moments (NEC) Cl] is one of those codes which
has entered this select category. There are
several reasons for this situation, including:
1) continuity of support and personnel involved
in its development; 2) systematic updating and
extension of its capabilities; 3) extensive,
user-oriented documentation; 4) its ready
availability; and 5) accessibility of its
developers for user assistance in applications. The result is that there are estimated
to be several hundred users of various versions
of NEC world wide.
NEC has been under development for more
than 10 years (in earlier forms it was known as
BRACT and AMP). It is a hybrid code which uses
an Electric Field Integral Equation (EFIE) to
model wire-like objects and a Magnetic Field
Integral Equation (MFIE) to model surface-like
objects with time harmonic excitation. A threeterm sinusoidal spline basis is employed for the
wire current while a pulse basis is used for the
surface current, with delta-function weights
employed everywhere. Provision is made for
connection of wires to surfaces via an
attachment, or interpolation, basis.
NEC includes a number of features for
efficient modeling of antennas and scatterers in
their environments. The most recent addition,
in the version NEC-3 [2,31, has been the
capability of modeling wires that are buried or
penetrate the ground-air interface. This paper
addresses the current status of NEC-3 and plans
for "Future NEC." Some typical results are
included.
THE SOLUTION METHOD
II.
Integral Equations
NEC-3 uses both an electric-field integral
equation (EFIE) and a magnetic-field integral
equation (MFIE) to model the electromagnetic
response of general structures. Each equation
has advantages for particular structure types.
The EFIE is well suited for thin-wire structures
of small or vanishing conductor volume while the
MFIE, which fails for the thin-wire case, is
more attractive for voluminous structures,
especially those having large smooth surfaces.
The EFIE can also be used to model surfaces and
is preferred for thin structures where there is
little separation between a front and back
surface. Although the EFIE is specialized to
thin wires in this program, it has been used to
represent surfaces by wire grids with reasonable
success for far-field quantities but with
variable accuracy for surface fields. For a
structure containing both wires and surfaces the
EFIE and MFIE are coupled. This combination of
the EFIE and MFIE was proposed and used by
Albertsen, Hansen, and Jensen at the Technical
University of Denmark [4] although the details
of their numerical solution differ from those in
NEC. A rigorous derivation of the EFIE and MFIE
used in NEC is given by Poggio and Miller [5].
The thin wire approximation is applied to
the EFIE to reduce it to a scalar integral
equation. The assumptions involved are:
a.
b.
c.
d.
Transverse currents can be neglected
relative to axial currents on the
wire.
The circumferential variation in the
axial current can be neglected.
The current can be represented by a
filament on the wire axis.
The boundary condition on the electric
field need be enforced in the axial
direction only.
These widely used approximations are valid
as long as the wire radius is much less than the
wavelength and much less than the wire length.
*Work performed under the auspices of the U.S. Department of Energy by the
Lawrence Livermore National Laboratory under contract No W-7405-ENC-48.
-
302
An alternate kernel for the EFIE, based on an
extended thin-wire approximation in which
condition c is relaxed, is also included in NEC
for wires having too large a radius for the
thin-wire approximation [6]. With the thin-wire
approximation the EFIE becomes
-
for F on wires and
i,(F)
l
i?(F) - - &
t,(F)
I
l
I(s’)(E
x 0’
L
-3
2(F) = 2
l
I(s’)(k2P
.
9’
g(;,;‘))ds’
-
- 3
t*,(F)
l
Is(G)
-
C
~1
a2
PC:,
;
i’)ds’
E
Z(s)
1
ZG f
:2(G)
l
[zs(:‘)
X V’
g(;,;‘)]dA’
(2)
s1
where
and
g(;, ;') = exp(-jk(r,
;‘I)/\;
-
;‘I
-i,(i)
k =WV’!J E
00
l
i+(r) = & f,(T)
l
I
I&')(2
x 0'
L
g(t,;‘)ds’
I is the induced current, E' is the exciting
field, s^ and s^'are unit vectors tangent to the
wire at s and s', and i;and P' are vectors to
the points s and s' on the wire contour C.
x
g(F) =:- 3 3p>
1
G
n(F)
+
x V’
ES
where t?(P) is $e unit vector normal to the
surface at P, H is the exciting magnetic field,
and j is the surface current. This equation is
separated into the coupled scalar integral
equations for components of the surface current.
For a structure with both wires and
surfaces, the hybrid EFIE and MFIE equations are
l
8’
V' g(r,?)ldA'
(3)
The integral equations (l), (Z), and (3)
are solved numerically in NEC by a form of the
method of moments. The test functions for the
method of moments solution on both wires and
surfaces are chosen to be delta functions
resulting in a point sampling of the fields
known as the collocation method of solution.
Wires are divided into short straight segments
with a sample point at the center of each
segment, while surfaces are approximated by a
set of flat patches or facets with a sample
point at the center of each patch. Straight
segments and flat patches are not mandatory for
the solution method but are used to simplify the
specification of the geometry and evaluation of
the fields.
Delta functions are also used in NEC as the
current expansion functions on surfaces. For
the MFIE this elementary Galerkin's method has
been found to provide good accuracy on large
smooth surfaces. Due to the nature of the
integral-equation kernels, however, the choice
of current expansion functions is more critical
in the EFIE than in the MFIE.
a2
asasl)
- -
L
g(;,;‘)dA’
f i,(i). [?$l)x
+
Numerical Solution
g(;,;‘)]dA’
S
I(s’)(k2$
9 js(G)
for F on surfaces. The vectors i,(F) and t"*(P)
are orthogonal unit vectors tangent to the
surface at F, and the symbol / represents
integration over wires while,
represents
integration over surfaces exclu
+ Ing wires.
X [j,(?‘)
;
S,(G)
s1
The MFIE for a closed surface s is
a(;)
1
z?
- 5
(1
The expansion functions for the current on
wires are chosen so that the total current on
segment number j has the form
Ij(s)
= A. + Bjsin
3
k(s-sj)
Is-ss(
+ Cjcos
kb-sj)
< Ai/2
J
(4)
J
56
303 -
where s. is the value of s at the center of
SegmentJj and A. is the length of segment j.
This expansion das first used by Yeh and Mei L-71
and has been shown to provide rapid solution
convergence [S], [9]. It has the added
advantage that the fields of the sinusoidal
_
currents are easily evaluated in closed torm.
The amplitudes of the constant, sine, and cosine
terms are related such that their sum satisfies
physical conditions on the local behavior of
current and charge at the segment ends. On a
single wire with continuous radius, obvious
conditions are that the current and charge
density (aI/as) are continuous along the wire
and that the current goes to zero at free wire
ends. These conditions applied at segment ends
together with the equations from the method of
moments are sufficient to determine the current
in the form of (4) on all segments.
If the wire radius is discontinuous, Wu and
King Cl01 have shown that the charge density
should satisfy the condition
Jl
Kirchhoff's law and (5) for each segment at the
junction. If an end of segment j has no
connections, then f. goes to zero there with no
condition on the de&vative.
The general
equations for fj(S) are given in Cll, Part I.
Where a wire connects to a surface, a more
realistic representation of the surface current
is needed than the delta function expansion
used in NEC is
normally used. The treatment
quite similar to that used by Albertsen et al.
c41. In the region of the wire connection, the
surface current contains a singular component
due to the current flowing from the wire onto
the surface. The total current on the four
patches about the connection point, with
coordinates shown in Fig. 1, is represented as
4
S,(X,Y)
= Iofbw’)
+
1 gj(x,y)
(sj-Iosj)
(6)
j=l
where I, is the current at the base of the wire,
and
x2+ yy
h
P(x,y)
=
2?T(x2 + y2)
=-.-L-L_
s at
junction
q$)
- y
(5)
5 =3(x,y)
j
s j
j
where a = wire radius,
k = 21~/X
y = 0.5772 (Euler's constant).
Q is related to the total charge in the
vicinity of the junction and is constant for all
wires at the junction.
,At a junction of several wires (5) is
applied on each wire. The continuity of current
is generalized to Kirchhoff's current law and
provides an equation to eliminate the unknown Q.
qx%Y)
= 1
(d+x)(d+y)
4d2
.!J2(X’Y)
= -+
(d-x) (d+y)
4d
.qX’Y)
=
1
(d-x) (d-y)
4d2
g4(x’Y)
=
L
(d+x) (d-y)
4d2
To enforce these conditions on total
current, the current on a structure with N
segments is expanded as
I(s)
1
=
j&jfj(5)
with the expansion functions fj(S) chosen to
satisfy the above conditions. On a uniform wire
f.(s) is a spline-like function extending over
s4 gments j-l, j, and j+l.
In general, if
several wires connect to each end of segment j,
f.(s) extends over segment j and all connected
sigments and has the form of (4) on each
segment. The constants A, 8, and C differ on
each segment and are chosen so that f. goes to
zero with zero derivative at the oute$ end of
each segment connected to segment j. At the
junctions at each end of segment j, fj satisfies
Figure 1. Detail of the connection of a wire
to a surface.
-
and (xj,yj) =
(x,Y)
304
at the center of patch j.
The current in (6) is integrated
numerically when computing the electric field at
the center of the connected wire segment due to
the surface current on the four surrounding
patches. In computing the field on any other
segments or on any patches, the delta-function
form is used for all patches including those at
the connection point. This saves integration
time and is sufficiently accurate for the
greater source to observation-point separations
involved.
ground surface. The solution is based on the
Sommerfeld-integral formulation for the field
near the interface. For the method-of-moments
solution, the field values are obtained by table
lookup and parameter estimation involving a
model for the complex behavior of the field
transmitted across the interface. The current
expansion is modified to account for the
discontinuity in charge on a wire penetrating
the interface. This numerical treatment for
burial wires is described in [2].
Efficient Solution Methods
III,
CAPABILITIES
OF NEC-3
NEC-3 includes a number of features for
convenient modeling of antennas and
scatterers. These are summarized below.
Source Modeling
A voltage source on a wire may be modeled
by an applied field on a segment or a
discontinuity in charge between segments.
Alternatively, a structure may be excited by a
plane wave with linear or elliptic polarization
or by the near field of an infinitesimal current
element.
Nonradiating Networks and Transmission Lines
Nonradiating two port networks and
transmission lines may connect points on
wires. These are modeled by deriving a drivingpoint admittance matrix from the admittance
matrix of the entire structure to avoid
modifying the larger matrix.
Loading
Lumped or distributed RLC loads may be
specified on wires. Also, the conductivity of a
round wire may be specified and the impedance
computed taking account of skin depth.
Ground Effects
Three options are available for an antenna
in or near the ground. A perfectly conducting
ground is modeled by including the image field
in the kernel of the integral equation. This
doubles the time to compute the interaction
matrix. An approximate model for a finitely
conducting ground uses the image modified by
Fresnel reflection coefficients. This
approximation is usable for antennas at least
0.1 to 0.2 wavelengths above the ground and
doubles the time to fill the matrix.
NW-3 includes an accurate treatment for
wire structures above, below, or penetrating the
The matrix equation is solved by factoring
the matrix into upper and lower triangular
matrices which are saved for reuse when the
excitation or other parameters that do not alter
the matrix are changed. Rotational symmetry or
reflection symmetry in one to three planes can
be used to reduce both the times to fill and
factor the matrix. New wires or surfaces may be
added to a structure for which the matrix has
already been computed, factored and saved on a
file. The new solution is found from the self
and mutual interaction matrices through a
partitioned-matrix algorithm with no unnecessary
repetition of calculations for the basic
structure. This feature can be used to take
advantage of symmetry in a portion of a
structure to which unsymmetric parts connect.
Input
A user-oriented input scheme permits
defining straight wires, arcs, and surfaces.
Electrical connections are determined in the
program by searching for wire ends and patch
centers that coincide. Shifts, rotations, and
reflections can be used in building complex
structures. A solution can be repeated for
modified model parameters (transmission lines,
loading, etc.) without respecifying parameters
that are not changed.
output
Output selectable by input parameters may
include:
current
charge density on wires
input impedance, admittance, and power
radiated power, ohmic loss, efficiency
radia,tedfield, power gain, directive
gain
average gain_
near E and H fields
maximum coupling (for matched source
and load)
receiving patterns
scattering cross section
For accurate results, the lengths of wire
segments should be less than about 0.1 A.
Longer segments up to about 0.15 x may be
acceptable on long, straight wires or
-
305
The wire radius, a, rf!lat.iVe
t0
X is
by the approximations used in the kernel
of the electric field integral equation. Two
approximation
options are available in NEC: the
thin-wire kernel and the extended thin-wire
kernel. In the thin-wire kernel, the current on
the surface of a segment is reduced to a
filament of current on the segment axis. In the
extended thin-wire kernel, a CUrtWIt
Uniformly
distributed around the segment surface is
assumed. The field of this current is apprOXimated
by the first two terms in a Series
expansion of the exact field in powers of a*.
With the extended thin-wire kernel, the
ratio of segment length to radius can be as
small as 0.5 before instabilities appear in the
solution, while with the thin-wire
approximation, the limit is 2. In either of
these approximations, only currents in the axial
direction on a segment are considered, and there
is no allowance for variation of the current
around the wire circumference. The acceptability of these approximations depends on both the
value of a/x and the tendency of the excitation
to produce circumferential current or current
variation.
Validation of results is an important step
in the modeling process. The ideal validation
is comparison with reliable measured data for
the antenna of interest. Lacking such data
there are several checks that can be made on the
internal consistency of the NEC solution. One
is to vary the density of segments or patches to
test the convergence of the solution. Since the
solution for wires uses different functions for
current expansion and field testing, reciprocity
is not assured. Hence, it is useful to check
reciprocity in the receiving and transmitting
patterns or bistatic scattering. The accuracy
Of the computed input resistance can be checked
(except for
antennas
over lossy ground) by
integrating the power in the radiated field.
The power obtained by integrating the radiated
field is insensitive to small errors in the
computed current and can, if necessary, be used
to correct the computed radiation resistance,
input power and values of gain.
An important consideration in using NEC is
the solution time versus model size since this
may limit the amount of detail that can be
modeled and the segment and patch densities.
For a model using N wire segments, the solution
time in seconds on a CDC 7600 computer is given
approximately by the formula
T = 3(10-4>kN2/M -I-2(10-6)~3/~2
where
and
(7)
M is the number of degrees of symmetry
k = 1 for free space
= 2 for perfectly conducting ground
Jl
or reflection coefficient
approximation
noncritical
parts of a structure while shorter
segments,
0.05 A or less may be needed in
modeling critical regions of an antenna.
A
reasonable
maximum for the area of a surface
patch appears to be 0.04 square wavelengths.
limited
56
-
4 to 8 for accurate (Sommerfeld)
tr:atment of finitely
conducting ground.
The first term in (7) is the time to fill the
matrix and the second term is the time to factor
it into triangular parts for solution. For a
model using No patches, the solution time is
I_
about
T = (10-5)k(2Np)2/M
+ 2(10-5
(2NJ3/M2
since two rows and columns in the matrix are
associated with each patch. The reduced time
factor for filling the matrix, due to the deltafunction current expansion, is one attractive
feature of this solution method. A more
complete relation for running time is contained
in [ll, Part III.
When the Sommerfeld ground treatment is
used, a fixed time of about 15 seconds on a CDC
7600 computer is needed to generate the
interpolation tables. The tables depend only on
the ground parameters and frequency, however,
and can be saved for use in any case in which
these parameters are the same.
IV.
REPRESENTATIVE RESULTS
NEC has been used to model a wide variety
of structures including LP and Yagi-Uda arrays;
Beverage, Discone-Cage and multiarm helical
antennas; spheres, cylinders, and cone-sphere
scatterers. Whip antennas mounted on a ship or
a simple box have been modeled using both the
wire-grid and patch methods.
Some results obtained with NEC-3 are shown
in Figs. 2, 3, and 4. The radiation pattern of
a cylinder with two attached wires is shown in
Fig. 2. The cylinder height is 22 cm, the
diameter 20 cm, and the wire dimensions are
shown in the figure. Wire 'a' was driven by a
voltage source at its base, while wire 'b' was
connected directly to the cylinder. This model
used the hybrid EFIE-MFIE capability of NEC.
The measured pattern was obtained by Albertsen
et al. C4l who presented similar numerical
results. NEC results for a higher density of
patches than was used for Fig. 2 were in worse
agreement with the measurements, possibly due to
inability of the MFIE to handle the square edges
of the cylinder.
The current on a monopole and radial-wire
ground screen is shown in Fig. 3 for the screen
above and below the ground surface. The sum of
the currents along the radials is plotted for
negative distance, while positive distance is
along the monopole. The transition from upper
to lower medium propagation factor for current
along the radials is seen-go take place within a
vertical distance of 2(10 )ho.
-
ol0
306
1
-
I
. 270
I
90
180
J
360
OBSERVATION ANGLE
Figure 2. Radiation pattern of a cylinder with
attached wires, with wire a excited.
In Fig. 4, a buried horizontal pipe is
illuminated by the field of a distant vertical
tower. The plot shays the magnitude of the
scattered field E (4 is normal to the direction
to the transmittifig tower) relative to the
incident field E, and represents measurements
made by moving a probe in a raster-like fashion
over the search area. Both active and passive
geophysical searches have been modeled with NEC3.
V.
-16. ’
-.5
’
’
-.4
n
-.3
-.2
Distance
-.l
0.
.1
.2
(Wavelengths)
Figure 3. Real part of the current on a quarter
wave monopole driven against a ground screen
consisting of six evenly spaced radial wires
with a screen radius of 0.5 h0 and 2_ = 16 j0. The height of the ground screen above the
interface is s. The sum of the radial currents
is plotted to obtain a continuous current at the
monopole-screen junction. It can be seen that
transition from the upper to lower medium
wavelength occurs6largely within a vertical
distance of 2(10 )ho.
CONCLUSION
NEC-3 is a versatile code for modeling
antennas and their environment including
transmission lines, networks, loading, and
ground effects. It has become widely used, due
mainly to its convenient operation, documentation, availability of the code, and continuing
support. NEC is written in Fortran IV, consists
of about 9,000 lines of code and requires about
95,000 words of memory on a CDC 7600. These are
also versions of the code on IBM 3033 and DEC
VAX-11 computers.
In a project now beginning, NEC is to be
revised with the new code to be called Numerical
Electromagnetics Engineering Design System
(NEEDS). In NEEDS, we hope to develop a truly
user-friendly code, taking advantage of both
user experience with the present code, and new
concepts in software design. One goal is to reengineer NEC into a tool that is more transportable. maintainable. and uodatable bv utilizina a
56
Jl
Buried Pipe
Figure 4. Typical signature (defined by the
magnitude of E relative to E ) of a buried
horizontal pip& (length L = 56 m, diameter d =
.2 m) buried 5 m below the interface for Z_ = 16
- j80 and f = 200 kHz. The excitation is
provided by a vertical electric source 10 km
distant. Only the left half of the field, which
is left-right symmetric, is plotted. The peaks
are associated with the charge accumulation on
each end of the pipe.
more module structure. The other primary goal
is to incorporate features that make NEC easier,
and therefore more productive to use. This will
be done through introduction of more diagnostics
and error checks, more use of standard geometry
modules, incorporation of interactive graphics,
etc.
In addition, a number of new modeling
capabilities are planned for NEEDS, including:
0
l
l
l
Electric field integral equation for
surfaces with the capability of
connecting wires to surfaces.
Coupling to GTD codes for reflector
antennas and scattering by plates and
cylinders.
Improved numerical precision on 32-bit
computers.
Modeling insulated wires in the ground
or air.
l
l
l
l
Propagation calculations for irregular
ground.
Further exploitation of symmetries and
repetition patterns to reduce solution
time.
Improved model for voltage sources.
Built in optimization capability for
design applications.
Plans for NEEDS were developed, in part, .
through a survey of the needs of current NEC
users. It is hoped that this interchange
between NEC users and developers and among users
can be continued through a NEC/NEEDS User's News
Letter published periodically. The news letter
would provide a forum for exchange of user
information gained in practical applications as
well as dissemination of information on new
developments in the code. Through this process,
we hope to develop an expanding set of modeling
guidance in a variety of applications.
- 308
-
REFERENCES
Cl1
Burke, G. J. and Poggio, A. J., "Numerical
Electromagnetic Code (NEC) - Method of
Moments Parts I, II and III," NOSC TD 116,
Naval Ocean Systems Center, San Diego, CA,
18 July 1977 (NEC-1) revised 2 January
1980 (NEC-2).
C81
Neureuther, A. R. et al., "A Comparison of
Numerical Methods for Thin Wire Antennas,"
presented at the 1968 Fall URSI Meeting,
Dept. of Electrical Engineering and
Computer Sciences, University of
California, Berkeley, 1968.
c21
Burke, G. J. and Miller, E. K., "Modeling
Antennas Near to and Penetrating a Lossy
Interface,' IEEE Transactions on Antennas
and Propagation, Vol. AP-32, No. 10, pp.
1040-1049, October 1984.
cg1
c31
Burke, G. J., "Numerical Electromaqnetics
Code - Method of Moments User's Guide
Supplement for NEC-3 for Modeling Buried
Wires," Lawrence Livermore National
Laboratory, UCID-19918, October 1983.
Miller, E. K., R. M. Bevensee, A. J.
Poggio, R. Adams, and F. J. Deadrick, "An
Evaluation of Computer Programs Using
Integral Equations for the Electromagnetic
Analysis of Thin Wire Structures," UCRL75566, Lawrence Livermore National
Laboratory, CA, March 1974.
DOI
Wu, T. T. and King, R. W. P., "The Tapered
Antenna and Its Application to the
Junction Problem for Thin Wires," IEEE
Transactions on Antennas and Propason,
Vol. AP-24, No. 1, pp. 42-45, January
1976.
c41
Albertsen, N. C., Hansen, J. E., and
Jensen, N. E., "Computation of Spacecraft
Antenna Radiation Patterns," The Technical
University of Denmark, June 1972.
c51
Poggio, A. J. and Miller, E. K., "Integral
Equation Solutions of Three-Dimensional
Scatterinq Problems," Chapt. IV in
Computer Techniques-for Electromagnetics,
edited by R. Mittra, Pergamon Press, NY,
1973.
Ccl
Poggio, A. J. and Adams, R. W.,
"Approximations for Terms Related to the
Kernel in Thin-Wire Integral Equations,"
UCRL-51985, Lawrence Livermore National
Laboratory, CA, December 19, 1975.
‘This document was prepared as sn account of work \penwrrd
the I’nitcd
Stales (;evernment.
Wversity
of (‘alifornia
press or implied,
curary.
Neither
hy sn agency of
the (‘nited State\ Govcrnmmt
or awumcs any legnl liahilily
complctenesc. or uwfulncw
or respwnihilily
of any inform&m,
Yeh, Y. S. and Mei, K. K., "Theory of
Conical Equiangular Spiral Antennas," Part
I - Numerical Techniques, IEEE
Transactions on Antennas amropagation,
AVol. P- , p.
ex-
for the w-
appsralw.
preducl, or
process disclosed, or rcprewnts thrl its USCwould not infringe privately owned
rights. Referenre herein tu any specific commercial productr, procre.
c71
nor the
nor any of their empleyecb, makes any warranty.
hy trade name, trademark.
manufacturer.
constitute or imply its endorwmcnl,
or otherwise.
rrcommcndatien,
Stater Government or the University
of ~‘alifornia.
or wnice
doe\ no1 ncwwrily
or fweriog
hy Ihe I’nitsd
.I'he
views and opinion% or
awlhers expreasad herein de not necessarily Qrte or rellecl thow of the I’niled
States G~~~rnent
dorwmeot
thereof, and shall not IJP uwd for adwrti+ing
purposes.
or product cn-
-
CClvB?UTER-AIDED ANALYSIS
IN VHF-FM
309
57~2
-
OF ELECTROMAGNETIC
BROADCASTING
D. J. Bern, J. Janiszewski,
Technical
University
Wrockaw,
The paper
method
presents
a computer-aided
compatibi-
for electromagnetic
lity analysis
casting
(EMCA)
The method
networks.
developed
in VHF-FM
on the basis
mathematical
model
broad-
has been
of an original
of the VHF-FM
net-
NETWORKS
R. Zielidski
of Wroclaw
Poland
The coverage
The protection
considered,
is determined
called
compatible
mitters
which
compatible
tic environment
dition
requires
of compatibility
all systems
information
should
compatibility
of various
signals.
tions provide
kinds
Simulation
us with
the performance
making
external
interferences.
number
ters distributed
frequency
achievable
ceivers,
with
a
and, at
from
co-
transmit-
predetermined
method
presents
a computer-aided
for electromagnetic
compatibi-
(EMCA) in
networks.
VHF-FM
The computer
in
FORTRAN
broadprogram
1900.
munication
ponents
mitter
consists
of
transmit-
should be
area.
of
done in
area,
i.e.,
a good reception
domestic
of the system elements
In a computer
of
and allocation
normal
value)
adjacent-channel
Models
for possible
such a way that the service
the region where
in the
ever a definite
channels
the field
of distur-
of radio
This distribution
and
casting
and analysing
network
(protected
ters do not exceed
investiga-
allowances
A broadcasting
value
has been written
broadcasting
networks
which
is not less than the minimum
lity analysis
this
characteristics
and existing
a certain
In the
an effective
for forecasting
planned
signal
area. The
range of a transmitter
The paper
an undisturbed
of a desired
presence
method
network
so
of the trans-
this
the same time, interference
channel
the
level.
of
be observed.
means
to
signals
or processing
case of a broadcasting
bing
regard
using electrical
for transmission
reception
that the conwith
by
ranges
service
an area over
usable
of the electromagne-
i.e., the ratio Of
area to the total area
strength
Introduction
ratio,
the service
means
work.
COMPATIBILITY
is
re-
model
system
of a radiocom-
the following
can be distinguished:
models,
antenna
com-
trans-
models,
recei-
ver models,
propagation
models,
and a
geographical
environment
model.
These
models
describe
system
components
all properties
in compatibility
Propagation
which
are
of the
important
analysis.
model.
The propagation
curves based
370-4
on CCIR
Recommendation
are used for calculations of the
field strength.
strength
to path
covers
the greatest
poss'ible
valent
Part of a given
geographical
area.
a parameter
They relate
length with
transmitting
antenna
for various
field
the equiheight
percentages
as
of
- 310 -
in various
time from 50% to 1%
They represent
tic regions.
strength
exceeded
and apply
model
for
vegetation
Pr = -119.6
allowance
and to the limit value
and man-made
structures
surface.
paths
and
However,
terrain
the territory
cluded
is enough
the transmitter
follows
analysis
in-
patibility
the considered
for
emissions
as a determined
Power
nominal
distribution
spectrum
com-
showed
are beyond
range.
a frequency-dependent
ratio
The receiving
characterized
antenna.
(ij protected
by
part of the system
- rural
is
para-
of the field
emission:
areas, 6Q dBp - urban
70 dBp - great
emission:
- urban
value
(monophonic
cities;
54 dBp - rural
areas,
(ii) noise
mismatch
transmitting
and receiving
the omnidirectional
mismatch
power
are
not taken
adopted
model
value of the
the reception
absence
is correct
of interference
ble field strength).
value
66 dBp
cities);
field
prefer-
Tab.
1. The
antenna
since
in
is
the
it does
Table
in the
Transmitting
antenna
vertical
horizontal
circular
linear
Attenuation
coefficient
Receiving
antenna
horizontal
vertical
linear
circular
for the effective
ing conditions
sary
height
equal
to
e.g., when
to know
smaller
represented
3
made
of ,the recei-
10 m.
in cars,
In some
the
it
receiv-
.is neces-
field strength
and then
for
an additional
is introduced.
by
antenna
can
a probabilistic
in the case of an
nal antenna
18
testing
the
heigths
correction
CdB]
curves were
\
be
model
omnidirectio-
takes the form
z
usa-
1
Signal impairment due to
polarization
mismatch
which
for
(minimum
antennae
of EMC analysis
The transmitting
is the minimum
of
the
is
in
into account
cases,
level at the receiver
The protected
between
presented
method
areas,
input.
strength
case
gain of the receiving
ving antenna
stereophonic
74 dBp - great
the
polarization
48 dBp
areas,
of an omnidi-
In
Polarization
[2].
by two additional
recommended
or the model
rectional
CCIR propagation
strength
which
[4]
antenna
to
meters:
._
by
in the transmit-
introducing
protection
a directional
and the receiver
are taken into account
According
of
power[l] .
selectivity
models.
to the needs one can use the model
is
equal
output
is described
not play any role.
The
emission
value
antenna
red. The consequences of polarization
of the
to the internal
of the fundamental
ter output
The
Cl1f
emission
frequency
the transmitter
It
takes into account
of the system which
that the harmonic
treated
model.
from the character
related
of the nuisance
models
by two deterministic
CCIR
only the fundamental
power
has been
and the receiver.
model
dBm
E = -1.6 dBu
The receiving
deover
to use a common model
transmitter
This
A
Ah,
irregularities
and receiver
[3)
Antenna
in the model.
Transmitter
susceptibi-
field strength
on
irregularities.
of Poland,
cor-
it takes
map of the parameter
scribing
by
lity threshold
land-sea
mixed
terrain
to the receiver
pro-
introduced
consideration
figure was aSSU-
to 4,5 dB which
and
does not make
into
responds
noise
The discussed
attenuation
the Earth's
digital
med to be equal
ver-
to both horizontal
field
The receiver
field
at 50% of locations,
tical polarization.
pagation
clima-
the
[l]
:
- 311
57~2
-
where e is the mean value of the power
gain in dB, or by means of a determini-
The other method assumes that there is
no time and spatial correlation
stic model in the form of a measured
radiation pattern.
between the fields and that the wanted
field strength is dominant everywhere.
The point here is to determine - for
EMC analysis
a given percentage of time -
The fundamental task of the electromagnetic analysis is to determine the
probability p that the wanted signal
level exceeds the level of a resultant
interfering signal by a strictly defined value over a given percentage
of time
P=
PC(EU - EI) > ACT,)]
(2)
where EU - wanted field strength in dB
EI - resultant interfering field
strength in dB
protection
ratio for Ta
ACT,) percent of time in dB.
EU is a two dimensional random
variable of Gaussian distribution
with median E(50,50) and standard
deviations UT, DL.
From the analysis of the CCIR propagation curves the standard deviation
of spatial distribution
UL = 6 + 0.69a
-
- O.O063(Ah/h)
and of time distribution
(3)
YT = 0.5I0.429 [E(1,50) - E(50,50)] +
+ 0.781[E(10,50) - E(50,50)])
(4)
were found.
tial probabilities of undisturbed reception in the presence of successive
L. . The probabi7
lity of undisturbed reception in the
presence of all interfering signals
is equal to the product of probabiliinterfering signals
ties for individual interfering signals
= II Lj
LE
j
The calculation of the
omnidirectional antenna
the standard deviation of spatial
distribution of the field strength is
calculated from the formula
(5)
is the standard deviation
caiculateci from
formula (31.
There are two methods to determine
the probability distribution of the
random variable EI. The first, which
is more laborious and sometimes gives
no results, consists in defining the
distribution function as the convolution of successive interfering fields.
-
Lj probabi-
input:
usu
= HA + ESU(50,50) + PPATRU +
(7a)
+ 'EFHU + pPoLu
usI = HA + ESI(50,50) + PPATRI +
(7b)
+ 'EFHI + pPoLI
where U - signal median at the receiver
S
input (index U means wanted
signal and I - interfering
signal)
HA - effective lengthof the receiving antenna
ES(50,50)- median of field strength
'PATR
at the receiving antenna
- correction resulting from
the radiation pattern of
the receiving antenna
model of an
where 0L
(6)
lity starts with the determination of
the signal medians at the receiver
In the case of the probabilistic
u;,=&Y
the spa-
'EFH - correction due to the change
of receiving antenna height
pPoL - correction due to polarization mismatch
All quantities in eqs. (7a) and
(7b) are expressed in dB.
The median of the difference of the
wanted and interfering signals
AUs = Usu
-
USI = ESU - E
is
SI + Ap
(8)
where AP = (PpATRu - PpATRI) + (ppoLu- 'POLI)+ ('EFHU - 'EFHI) ('I
The difference in the levels of the
wanted and interfering signals exceeded
in T percent of time and in L percent
-'
- 312 -
is - according
of places
distribution
- described
AUS(L,T)
to the normal
which
by
last but not least,
(10)
= AUS - H(T) - H(L)
siderations.
H(T) = F(T) _I
0TU
(lib)
with
F(x) being
distribution
T
>
ratio depends
ference between
fering
of the protection
(11) to eq.
eqs.
and inter-
(7), (8)
(13)
ble
(14)
the spatial
lity of nondisturbed
The following
reception
parameters
in
EMC
probability
reception
[53:
of places
is fulfilled
with
Compatible
the value
exceeds
strength
included)
of the wanted
the minimum
signal
usable
segment
(MODY),
set of calculation
by the
as new transmitters
introduced
user may become
The
-
components
median
presence
of many
(in
point.
a
the
sources)
The SIP0
version
one; it selects
transmitters
to
EMC
performs
interfering
is a simplified
the previous
also the
of
analysis
observation
this
he wishes
procedure
where
at a given
which
well
by the
of
The user determines
POINT
net-
transmitter
of
only
at the observation
point.
with LVS
(16)
being
of the spatial
reception.
user of
takes into account
transmitter
Any
data file as
full interference
L* b LVS
limit value
who
from the permanent
useful
set
a particular
for analysis.
are possible.
(15)
is the
on disc,
one is the user
analysis
and the relation
of undisturbed
data file stored
Ta =
EU(5O,5O)>,EC
output
The first data source
permanent
Seven procedures
EC
algO-
of data are used in the
perform.
field
steering
particular
and data input and
rithms,
procedure
is fulfilled
realizing
kind of analysis
range of a transmitter
the area (boundaries
a supervisory
(Fig. 1)
where
99%.
in
systems.
data base with
network.
of undisturbed
- percentage
(2)
system
analysis
compatibility
MASTER,
selects
describing
was
segment
work
Lj.
program
1900 for the ana-
sound broadcasting
and the second
probabi-
sound broadcasting
are of interest
Spatial
VHF-FM
program.
the value H(L) one can now
determine
condition
is accepta-
=1-T
TX
computer
FORTRAN
segments.
TWO sources
of time
interference
VHF-FM
in
procedures
-
- A(T)
TX is the percentage
easily
and
- ESI(50,50)
+ AP + H(TX)
Knowing
NEWEMCA
described
of the
lysis of internal
(12) one gets
during which
method,
written
It contains
H(L)
gor 8'
We emphasize thatr, is a point estimator, embodying the specific structure, gE, (2.1), of the receiver performing the estimation. On the other
hand, interval estimators, as expressed
by P, (2.31, (2.4), yield a probability which is a measure of the efficiency of the point estimator for any particular application (i.e., choice of &).
The average error (or risk) (2.2) measures the expected cost or average error in using x0, considered over all
possible ($1 received.
For optimum, or Bayes estimation we
seek estimators YX which minimize the
average error or?isk R(o,6), (2.2).
The general form of the resulting y*
depends, of course, on the choice o B
"Cost“ or error function C&_ or Q,y,).
For example, for the quadratic cost
function (QCF)
c(~,&J)=colo-$J12 =
?
(2.5)
‘0 ,i, (em-Y$m)
Lf
the associated optimum estimator is
found to be (cf. Chapter 3, [51, Chap-
(2.6)
with A$, o(Q)+cr(s),a-+@ in the case
of estimating signal waveforms. Note
that 1% is generally a nonlinear operator on the received data, 8.
Another cost function of considerable interest is the simple (or rectangular) cost function (SCF) given by
M"
(2.7)
C(~,~c)=Co mil [Am'G(Ym-em)l
with appropriate choices of Co, Am to
ensure meaningful results (e.g., positive errors, etc.). Minimization of
average risk here leads directly to
(cf. Sec. 21.2-1 of [6]) the following
relations determining yg SCF:
I
a
log
ae
m
c4
0,)
w, (4 I.e,) 1
I
em=eg=y*
all m=l,...,M,
where WJ&l
=
0,
m
(2.8)
~,=e,)"
, and
)>p,=,_,_ sn /a(v,=e,)
a(Y,=f3,)=,, with Q'=all Q
except 8
The con& ition (2.8) deter92.
mining x={y&]
is precisely that determining the unconditional maximum like,=Sj_max
Accordingly we write
noise s2 mples,
Physically, our